Methods and apparatus based on coplanar striplines

ABSTRACT

A standing wave oscillator to generate at least one voltage standing wave, comprising a closed-loop coplanar stripline including two conductors, and at least one amplifier disposed between the two conductors at a first location. The two conductors are connected together at a second location different from the first location to provide a zero voltage node for the at least one voltage standing wave. One or more of a tailored distributed amplification scheme, a plurality of linear conductive strips disposed in proximity to the coplanar stripline, and a tapered coplanar stripline configuration may be used with the closed loop structure. A particular amplifier configuration involving cross-coupling of the coplanar stripline conductors may be employed to facilitate single mode operation, using a particular resonator topology so as to avoid inducing significant loss in the oscillator.

CROSS-REFERENCES TO RELATED APPLICATIONS

This application claims the benefit, under 35 U.S.C. §120, as adivisional of U.S. Non-provisional application Ser. No. 10/894,674,filed Jul. 19, 2004, and entitled “Methods and Apparatus Based onCoplanar Striplines.”

Ser. No. 10/894,674 in turn claims the benefit under 35 U.S.C. § 119(e)of U.S. Provisional Application Ser. No. 60/489,708, filed Jul. 23,2003, entitled “Methods and Apparatus for Implementing Standing WaveSinusoidal Oscillators,” and U.S. Provisional Application Ser. No.60/533,904, filed Jan. 2, 2004, entitled “Methods and Apparatus forImplementing Standing Wave Sinusoidal Oscillators.”

Each of the foregoing applications hereby is incorporated herein byreference.

FIELD OF THE INVENTION

The present invention relates generally to various methods and apparatusinvolving semiconductor devices based on coplanar striplines (CPS). Insome exemplary embodiments, sinusoidal signal sources, and moreparticularly standing wave sinusoidal oscillators, are implemented basedon coplanar stripline configurations.

BACKGROUND

Advanced communications applications that are now a routine part ofeveryday life, such as cellular telephones, wireless networks, satellitebroadcasting and fiber-optic communication, rely on continuing advancesin the electronic and related arts in connection with increased speedand reduced size; namely, increasing the speed of informationtransmission and miniaturizing the integrated circuits that performvarious communication-related functions. However, as system designerslook forward to using higher frequencies in the range of tens ofgigahertz (GHz), as well as the miniaturization of integrated circuitstoward an atomic scale, a number of aspects of conventional integratedcircuit technology continue to become inapplicable and obsolete. Thus, acommon design challenge involves finding new ways to implementwell-known circuit building blocks for faster operation, and often in asmaller space. In some instances, such implementations may exploitelectromagnetic wave-base concept, and involve a transmission line orwaveguide configuration fabricated on a semiconductor substrate.

Transmission line theory is well-established in the art. Generallyspeaking, a transmission line provides a means by which power orinformation may be transmitted in a guided manner, for example, toconnect a signal source to a load. A transmission line typicallyincludes two parallel conductors separated by a dielectric material.Signals propagate along a given transmission line as electromagneticwaves, and various physical parameters relating to the transmissionline, as well as parameters relating to the source of the signal on theline and any loads on the line, affect wave propagation.

FIGS. 1 a-1 e illustrate typical examples of transmission lines,including a coaxial cable (FIG. 1 a), a two-wire line (FIG. 1 b), aparallel-plate or planar line (FIG. 1 c), a wire above a conductingplane (FIG. 1 d), and a microstrip line (FIG. 1 e). Again, it isnoteworthy that each of these examples consists of two conductors inparallel. Coaxial cables routinely are used in electrical laboratoriesand in several common consumer applications to interconnect variouselectrical apparatus (e.g., connecting TV sets to TV antennas or cablefeeds). Microstrip lines are particularly important in integratedcircuits based on various semiconductor fabrication techniques, whereparallel metallic strips fabricated on a dielectric substrate (i.e.,separated by a dielectric) connect electronic elements.

Transmission lines often are regarded as special cases of a widercategory of “waveguides.” A waveguide refers generally to a system thatis configured to guide electromagnetic radiation from one point toanother. In several common applications, however, a waveguide is morespecifically regarded essentially as a bounded conduit through whichelectromagnetic radiation propagates in a somewhat more confined mannerthan that generally considered in connection with transmission lines.For example, in the microwave regime, unlike a two-conductortransmission line, a waveguide may be formed as a hollow metallic pipeor tube that may be rectangular, elliptical, or circular incross-section. In the optical regime, which is not supported at all bytransmission lines, waveguides often are formed as a solid dielectricfilament (e.g., an optical fiber) or a thin dielectric film bounded by alower refractive index environment.

As conventionally treated in many applications, transmission lines oftenmay be characterized somewhat differently from a wider category ofwaveguides in some significant respects. For example, first, atransmission line generally may be configured to operate from DC(frequency f=0) to very high frequencies (e.g., in the millimeter andmicrowave range, from about 1 GHz to 100 GHz); however, a waveguide canonly operate above a certain frequency (a “cutoff frequency”) determinedby its particular construction and dimensions, and therefore actsgenerally as a high-pass filter. On the other hand, at significantlyhigh frequencies on the order of approximately 50 GHz to 300 GHz,transmission lines conventionally are regarded as becoming generallyinefficient due to the well-known skin effect in the transmission lineconductors, as well as dielectric losses involving the materialseparating the conductors; in contrast, waveguides conventionally areconsidered in this range of frequencies to obtain larger bandwidth andlower signal attenuation (i.e., a wider range of frequency response withlower signal power loss). However, at the lower end of this frequencyrange and below, waveguides conventionally are considered to becomeexcessively large in size for some applications, especially integratedcircuit applications in which increased miniaturization typically is agoal. Yet another difference between transmission lines and waveguidesis that a transmission line can only support a transverseelectromagnetic (TEM) wave (i.e., a wave in which both the electric andmagnetic field are oriented transversely to the direction of wavepropagation), whereas a waveguide generally can support many possiblefield configurations (i.e. modes).

In semiconductor fabrication of microelectronic circuits, waveguides andtransmission lines for carrying high frequency electronic signalsconventionally have been implemented in a variety of ways. Two suchimplementations are referred to respectively as a coplanar waveguide(CPW) and a coplanar stripline (CPS). FIGS. 2A and 2B show differentviews of a coplanar waveguide, while FIGS. 3A and 3B show differentviews of a coplanar stripline.

In particular, FIG. 2A illustrates a cross-sectional view of a coplanarwaveguide 50 formed by three parallel conductors 20A, 40, and 20Bdisposed on a dielectric layer 101 on a semiconductor substrate 103.FIG. 2B shows a top view looking down onto an exemplary coplanarwaveguide device, in which the center conductor 40 is terminated on eachend with pads 42A and 42B, and the conductors 20A and 20B are shown tobe electrically connected so as to completely surround the conductor 40in the plane (the cross-sectional view of FIG. 2A is taken along thedashed lines 2A-2A in FIG. 2B). As illustrated in both FIGS. 2A and 2B,a width W₁ of the conductors 20A and 20B may be significantly greaterthan a width W₂ of the center conductor 40.

During normal operation, the conductors 20A and 20B of the coplanarwaveguide 50 are electrically connected together to a ground orreference potential, and the signal to be transmitted is applied to thecenter conductor 40. In this respect, it is particularly noteworthy thatthe respective ground and signal conductors in a coplanar waveguide arenot symmetric, as the combined ground conductors 20A and 20B cover asignificantly larger area than the center signal conductor 40. Thisconfiguration commonly is referred to as an “unbalanced” configuration.The arrangement of a large ground or reference potential surrounding thecenter signal conductor 40 serves to confine the electric field in theregions between the center conductor and the ground or referenceconductors, thereby creating the “conduit” through which the wave maypropagate.

In contrast to a coplanar waveguide, a coplanar stripline is a symmetricor balanced two-conductor device. FIGS. 3A and 3B show differentperspectives of one example of an idealized infinite coplanar stripline100 made up of two substantially identical parallel conductors 100A and100B separated by a distance S. In particular, FIG. 3A shows across-section of the conductors 100A and 100B which, for example, may bemetal lines disposed above the dielectric layer 101 on the substrate103. FIG. 3B shows a top view looking down onto the conductors disposedon the substrate (the cross-sectional view of FIG. 3A is taken along thedashed lines 3A-3A in FIG. 3B).

As can be readily observed in FIGS. 3A and 3B, the geometry of thecoplanar stripline 100 is significantly different from that of thecoplanar waveguide 50 shown in FIGS. 2A and 2B. In particular, thecoplanar waveguide 50 includes three conductors in cross-section,whereas the coplanar stripline 100 includes only two conductors.Moreover, unlike the ground conductors 20A and 20B and the center signalconductor 40 of the coplanar waveguide, which may have respectivelydifferent widths, the conductors 100A and 100B of the coplanar striplinehave substantially identical widths W₃, as indicated in FIGS. 3A and 3B.Again, this arrangement of substantially identical parallel conductorsin the coplanar stripline commonly is referred to as a symmetric or“balanced” configuration. Such a symmetric or balanced two-conductorconfiguration readily supports differential signals on the coplanarstripline, as discussed further below; in contrast, the asymmetric orunbalanced configuration of the coplanar waveguide does not supportdifferential signals, but merely supports “single-ended” signals (i.e.,a signal that is referenced to a ground potential).

For many conventional microwave applications, coplanar waveguideimplementations generally have been preferable as circuitinterconnection structures due to the prevalence of primarilysingled-ended or unbalanced microwave devices. Also, coplanar waveguidesgenerally have been regarded as significantly less lossy than coplanarstriplines, especially with respect to signal losses to the substrate atmicrowave frequencies. Hence, historically speaking, much of therelevant literature in connection with high frequency microelectronicdevices has focused significantly more on coplanar waveguides ratherthan on coplanar striplines. Coplanar waveguides generally are regardedas easily integrated with both series and shunt active and passivecircuit components. Moreover, the dimensions of coplanar waveguideconductors may be readily varied to match circuit component lead widthsand thereby facilitate interconnection with other devices, whilemaintaining a desired characteristic impedance for the coplanarwaveguide that is compatible with the interconnected devices. Onetradeoff, however, is that coplanar waveguides take up significant spacedue to the relatively wide and multiple ground conductors flanking thecenter signal conductor.

Various characteristics of both coplanar waveguides and coplanarstriplines may be modeled at least to some extent using common conceptsrelated to electric circuit theory, such as resistance, inductance,conductance and capacitance. Wave-based structures in general, however,differ from ordinary electric networks in one essential feature: namely,size relative to operating frequency. For example, whereas the physicaldimensions of electric networks are very much smaller than thewavelength corresponding to the operating frequency, the size of devicesbased on waveguides and transmission lines is usually a considerablefraction of the wavelength corresponding to the operating frequency ofthe device, and may even be many wavelengths long. Accordingly, whereaselements relating to resistance, inductance, conductance and capacitancemay be described in common electric circuits as discrete componentshaving lumped parameters, transmission lines and waveguides must insteadbe described by circuit parameters that are distributed throughout thelength of the transmission line/waveguide.

In view of the foregoing, FIGS. 4A and 4B illustrate two differenttheoretical transmission line/waveguide models involving distributed“line parameters” based on electric circuit concepts. In particular,FIG. 4A shows a “single-ended” model 30 (which may be applied to thecoplanar waveguide 50 illustrated in FIGS. 2A and 2B) and FIG. 4B showsa “differential” model 32 (which may be applied to the coplanarstripline 100 illustrated in FIGS. 3A and 3B).

In the models of FIGS. 4A and 4B, the parameter z indicates distancealong a length of the transmission line/waveguide in the direction ofwave propagation (where dz denotes differential length). Thecircuit-based line parameters are indicated in FIGS. 4A and 4B asresistance per unit length R, inductance per unit length L, conductanceper unit length G, and capacitance per unit length C, wherein R and Lare series elements and G and C are shunt elements. In FIG. 4B, thevalues attributed to the series elements R and L are divided amongst twoidentical conductors of the model 32 (e.g., Rdz/2 and Ldz/2), to againindicate the “differential” nature of the model.

The line parameters R, L, G and C that may be used to characterize acoplanar waveguide or coplanar stripline directly result from the typesof materials used to fabricate the coplanar stripline or coplanarwaveguide (e.g., the dielectric, substrate, and metal components) andthe various dimensions associated with the coplanar stripline orcoplanar waveguide arrangement (e.g., width and thickness of theconductors, space between the conductors, thickness of the dielectriclayer, etc.). More specifically, the materials and dimensions involvedin a given structure generally determine various physical propertiesassociated with the structure, such as effective permittivity ε_(eff),permeability μ, and various loss factors, on which the line parametersR, L, G and C are based.

Again, as illustrated in FIGS. 4A and 4B, it should be appreciated thatthe line parameters R, L, G and C are not discrete or lumped butuniformly distributed along the entire length of the coplanar striplineor coplanar waveguide. Also, it should be appreciated that R is the ACresistance per unit length of the conductors (i.e., “series”resistance), whereas G is the conductance per unit length due to thedielectric medium separating the conductors from each other and thesubstrate (i.e., “shunt” resistance).

The distributed resistance, conductance, inductance and capacitance ofthe coplanar stripline or coplanar waveguide naturally result inparticular frequency characteristics of a given implementation. Forexample, the general energy storage functions of both inductance andcapacitance have a frequency dependence based on anyresistance/conductance associated with the inductance/capacitance. Onecommon parameter for characterizing the frequency response of afrequency dependent system, including a given transmission line (orwaveguide) configuration, is referred to as a “quality factor,”typically denoted in the literature as Q.

The quality factor Q of a frequency dependent system generally isdefined as a ratio of a peak or resonant frequency of the system to thefrequency bandwidth of the system (i.e., the frequency range between thehalf-power points of the overall frequency response of the system). Thequality factor Q alternatively may be viewed as a ratio of the maximumenergy stored in the system to the total energy lost by the system in agiven time period. In view of the foregoing, systems with a relativelylarge Q generally are viewed as being “frequency selective,” in thatthey support frequencies close to a given resonant frequency withrelatively little energy loss. In contrast, systems with a relativelysmaller Q do not necessarily have a significant frequency preference andare often viewed as somewhat lossy systems.

The quality factor Q of a given coplanar waveguide or coplanar striplinearrangement also may be expressed in terms of various parametersassociated with the propagation of a wave along the coplanar waveguideor coplanar stripline. With reference again to the coplanar stripline100 shown in FIG. 3B, an exemplary position-dependent voltage V(z) isindicated between the conductors and an exemplary position-dependentcurrent I(z) is shown flowing through the conductors, where z indicatesdistance along the direction of wave propagation. The voltage V(z) as afunction of position z along the coplanar stripline may be expressed as:V(z)=V _(o) e ^(−az) cos(2πft−βz),where V_(o) is the amplitude of the wave, and the quantity (2πft−ρz)represents the phase (in radians) of the wave, which depends on bothtime t and space z. Of course, f is the frequency of the wave, and β isthe “phase constant” of the wave, defined as β=2π/λ; essentially, thephase constant β indicates that for every wavelength of distancetraveled, a wave undergoes a phase change of 2π radians. Finally, α isan attenuation factor representing losses as the wave propagates, whichaffect the overall amplitude of the wave; namely, as α increases,indicating greater loss, the amplitude V_(o) of the wave accordingly isdecreased by the factor e^(−az). As mentioned above, the quality factorQ also may be expressed in terms of the phase constant β and theattenuation factor α and approximated as Q≈β/2α for relatively low lossfrequency dependent systems.

Another important characterizing parameter of transmission lines andwaveguides relates to the speed with which waves propagate along thetransmission line or waveguide. In particular, the phase velocity of atransmission line or waveguide, commonly denoted as ν, provides therelationship between the frequency f and wavelength λ of a wave in agiven medium, according to ν=fλ, and represents the speed of wavepropagation in the medium. Accordingly, for a given frequency f, asmaller phase velocity ν results in a shorter wavelength λ. The phasevelocity ν results from the particular physical characteristics of thedevice, such as effective permittivity ε_(eff) and permeability μ. Withrespect to the models illustrated in FIGS. 4A and 4B, the phase velocitymay be expressed in terms of the inductance per unit length L and thecapacitance per unit length C as ν=1/√{square root over (LC)}.

Since reduction in circuit size generally is a significant goal ofimproved microelectronic device fabrication techniques, there has beenfocus in the literature in connection with size reduction of microwavedevices based on features that facilitate a reduction in phase velocity.Again, a reduction in phase velocity results in a correspondingreduction in wavelength at a given operating frequency. Devices such asresonators, oscillators, impedance matching networks, signal splittersand combiners, filters, amplifiers and delays may be implemented basedon transmission line or waveguide configurations. Often, as mentionedabove, the size of such devices is comparable with the wavelength λgiven a desired range of operating frequencies. Accordingly, by loweringthe phase velocity ν, smaller devices may be realized.

With this in mind, various “slow-wave” structures in the microwave fieldhave been investigated since the 1970s. Again, many of these studiesrelate to monolithic microwave integrated circuits (MMICs) involvingcoplanar waveguides that incorporate features designed to decrease thephase velocity and wavelength, and hence device size, at a givenoperating frequency or range of frequencies. One such feature forrealizing slow-wave structures includes a “periodically loaded” coplanarwaveguide, in which floating metal strips are placed periodicallybeneath the three coplanar waveguide conductors and orientedtransversely to the conductors. The presence of the floating metalstrips generally is considered to spatially separate the electric andmagnetic energy in the propagating wave, which results in an increasedcapacitance per unit length C of the coplanar waveguide. According tothe relationship ν=1/√{square root over (LC)}, such an increasedcapacitance per unit length C results in a smaller phase velocity ν, andhence a smaller wavelength λ at a given frequency f. Thus, the presenceof these slow-wave features may facilitate fabrication of smallerdevices.

In conventional slow-wave microwave structures based on coplanarwaveguides, a reduction in wavelength λ results in a correspondingincrease in the phase constant β, pursuant to the relationship β=2π/λ.However, the effect of increased phase constant β on the quality factorQ, according to the relationship Q≈β/2α, is not entirely clear from theliterature; while an increase in Q might be expected from an increase inβ, the effect of the slow-wave features on the loss α of the coplanarwaveguide is unclear. In some reports, it has been suggested that the Qof a coplanar waveguide slow-wave structure incorporating floating metalstrips actually may decrease from that of a coplanar waveguide withoutthe slow-wave features, due to increased loss α resulting from thepresence of the slow-wave features. Thus, it appears that there may be atradeoff between quality factor and phase velocity in some coplanarwaveguide slow-wave structures; namely, that while phase velocity may bereduced to facilitate implementation of smaller devices, greater lossesmay result, thereby degrading the quality factor Q of the device.

SUMMARY

The present disclosure relates generally to various methods andapparatus involving semiconductor devices based on coplanar striplines(CPS).

Although coplanar waveguides (CPW) have perhaps received greaterattention in the past in areas such as microwave circuit devices andstructures, Applicants have recognized and appreciated that variouscoplanar stripline (CPS) configurations may facilitate fabrication ofseveral useful high-speed microelectronic devices for a wide variety ofapplications.

Several differences between coplanar waveguides and coplanar striplineshave been discussed above in connection with FIGS. 2A, 2B, 3A and 3B.For example, coplanar striplines are two conductor arrangements incross-section, whereas coplanar waveguides are three conductorarrangements in cross-section, typically requiring significantly morespace than coplanar striplines. The two-conductor arrangement of acoplanar stripline is a “balanced” configuration due to the symmetry ofthe conductors; in contrast, coplanar waveguides are “unbalanced”configurations due to the significant asymmetry amongst the threewaveguide conductors (i.e., two wide conductors and one narrowconductor).

For many circuit applications, perhaps one of the most significantdifferences between coplanar striplines and coplanar waveguides is that,due primarily to their balanced configuration, coplanar striplines cansupport a differential signal, whereas a coplanar waveguide cannot.

Differential signals are important in applications where signals may beprone to contamination by “pickup” and other miscellaneous noise. Forexample, a signal transferred over relatively long distances, and/or inenvironments where several signals or other radiation may be present,may be subject to undesirable distortion that corrupts the integrity ofthe signal. By transporting a signal in a differential manner using twoconductors, any noise that is commonly picked up along both of theconductors may be cancelled out as the differential signal is recovered(by observing the difference between the respective signals on the twoconductors); specifically, the common-mode noise on the conductors is“rejected” by subtracting the signal on one conductor from the signal onthe other conductor, preferably leaving only the differential signal.This concept commonly is referred to as “common-mode rejection.”

The ability of a coplanar stripline to readily support differentialsignals and thereby facilitate common-mode rejection of undesirablenoise may be clearly observed with reference again to FIGS. 3A and 3B.In particular, in the coplanar stripline 100 shown in these figures,neither of the two virtually identical conductors 100A and 100B needs tobe at a signal ground or other reference potential; rather, both of thecoplanar stripline conductors may respectively and simultaneously carrydifferent signals that are each referenced to ground or some otherpotential. Furthermore, because the conductors are virtually identicaland proximate to each other, they respond essentially identically totheir environment in terms of noise pickup.

In contrast, a coplanar waveguide (as illustrated in FIGS. 2A and 2B)only supports a “single-sided” electrical signal, namely, a signal thatis referenced to a ground potential. Furthermore, the coplanar waveguideinherently is unbalanced based on the typically larger size of itscombined ground conductors compared to its signal conductor. Hence, theconductors of a coplanar waveguide respond differently to theirenvironment in terms of noise pickup. Accordingly, a coplanar waveguidedoes not readily support a differential signal, and devices based oncoplanar waveguides may not take advantage of the noise reductioncapability provided by coplanar striplines. Of course, a coplanarstripline also may be configured such that one of its two conductors isat a ground or some other reference potential; however, the capabilityof a coplanar stripline to support a differential signal makes acoplanar stripline configuration significantly more desirable than acoplanar waveguide configuration for many circuit applications.

In view of the foregoing, several embodiments disclosed below relate tocoplanar stripline configurations incorporating various features tofacilitate the implementation of a number of different microelectronicdevices. Examples of devices that may incorporate various coplanarstripline configurations according to the present disclosure include,but are not limited to, impedance matching devices, devices for powercombining and division, delays, resonators, oscillators, filters,amplifiers, mixers and the like, including CMOS-based implementations ofsuch devices. In some exemplary embodiments, sinusoidal signal sources,and more particularly standing wave sinusoidal oscillators, areimplemented based on various coplanar stripline configurations accordingto the present disclosure.

Some embodiments discussed further below relate to various features ofcoplanar stripline implementations that dramatically increase thequality factor Q of the resulting device. For example, in variousaspects of such embodiments, an enhancement of the quality factor Q onthe order of 20 or higher may be realized for coplanar stripline devicesfabricated on silicon substrates, as well as other substrates. Such anenhancement significantly and favorably contributes to improvedperformance of various circuit devices based on such implementations(e.g., resonators, oscillators). In one embodiment, an enhancement ofthe quality factor Q is achieved while at the same time reducing thephase velocity of one or more waves propagating in the device, therebyalso facilitating the fabrication of smaller devices. In yet anotherembodiment, a tapered coplanar stripline configuration results inposition-dependent line parameters, which may be exploited to achievesignificantly high-Q devices.

For example, one embodiment of the present invention is directed to anapparatus, comprising a coplanar stripline (CPS) including only a firstconductor and a second conductor essentially parallel to each other andoriented substantially along a first direction. The apparatus of thisembodiment further comprises a plurality of linear conductive stripsdisposed in proximity to the coplanar stripline. The plurality of linearconductive strips are essentially parallel to each other and orientedsubstantially along a second direction transverse to the firstdirection. In one aspect of this embodiment, the apparatus furthercomprises a silicon substrate on which the at least one dielectricmaterial, the plurality of linear conductive strips, and the coplanarstripline are disposed. In another aspect, the apparatus is configuredto support at least one signal on the coplanar stripline having afrequency in a range of from approximately 1 Gigahertz to 60 Gigahertzor higher. In yet another aspect, the coplanar stripline and theplurality of linear conductive strips are arranged such that theapparatus has a quality factor Q of at least 30 for at least onefrequency in the range of from approximately 1 Gigahertz to 60Gigahertz.

Another embodiment of the invention is directed to a method oftransporting at least one differential signal, comprising an act oftransporting the at least one differential signal over a coplanarstripline oriented substantially along a first direction and disposed inproximity to a plurality of linear conductive strips, wherein theplurality of linear conductive strips are essentially parallel to eachother and oriented substantially along a second direction transverse tothe first direction.

Yet another embodiment of the present invention is directed to acoplanar stripline configured such that the resistance per unit length Rand the conductance per unit length G are discreet or continuousfunctions of position along the coplanar stripline. In one aspect ofthis embodiment, a tapered coplanar stripline configuration isimplemented wherein a space between the coplanar stripline conductorsand a width of the conductors themselves is varied along a length of thecoplanar stripline. In one aspect of this embodiment, such a taperedconfiguration effectively changes the line parameters R and G along thelength of the coplanar stripline while substantially maintaining auniform characteristic impedance of the coplanar stripline, so as toavoid local reflections.

Another embodiment of the present invention is directed to an apparatus,comprising a tapered coplanar stripline including a first conductor anda second conductor, wherein the first and second conductors are orientedsubstantially along a first direction, and wherein a space between thefirst and second conductors and a width of the first and secondconductors is varied along a length of the coplanar stripline. Theapparatus of this embodiment further comprises a plurality of linearconductive strips disposed in proximity to the tapered coplanarstripline. The plurality of linear conductive strips are essentiallyparallel to each other and oriented substantially along a seconddirection transverse to the first direction.

Other embodiments of the present invention are directed generally tovarious methods and apparatus for implementing standing wave sinusoidaloscillators based on coplanar striplines.

For example, one embodiment of the invention is directed to aquarter-wavelength (λ/4) coplanar stripline standing wave oscillator(SWO) configured to generate at least one voltage standing wave having afrequency f_(o). The SWO of this embodiment comprises a coplanarstripline including two conductors and having a length L equal to orapproximately equal to a quarter-wavelength (λ/4), wherein λ is relatedto the frequency f_(o) by a phase velocity of waves constituting the atleast one voltage standing wave. The SWO further comprises at least oneamplifier disposed between the conductors at a first end of the coplanarstripline, wherein the two conductors are connected together at a secondend of the coplanar stripline to form a short circuit.

In one aspect of this embodiment, the SWO is configured to optimizesinusoidal performance at high frequencies with low power dissipation byemploying mode control techniques. In particular, in one aspect of thisembodiment, the SWO is configured as an essentially single mode deviceusing a tailored distributed amplification scheme employing multipleamplifiers having different gains along the length of the coplanarstripline. In yet another aspect of this embodiment, the different gainsof the amplifiers are “amplitude-dependent” in that they are based atleast in part on the expected amplitudes of the desired mode at therespective positions of the amplifiers along the coplanar stripline.

More generally, one embodiment of the invention is directed to a methodfor generating at least one voltage standing wave on a coplanarstripline, comprising an act of distributing amplification in a varyingmanner along the coplanar stripline so as to overcome coplanar striplineloss. Another embodiment of the invention is directed to a method forgenerating at least one voltage standing wave on a coplanar stripline,comprising an act of controlling an oscillation mode of the at least onevoltage standing wave. In various aspects of these embodiments,amplitude-dependent distributed amplification may be employed tofacilitate low loss single mode operation.

Another embodiment of the invention is directed to an SWO employing acoplanar stripline configuration comprising a plurality of linearconductive strips disposed in proximity to the coplanar stripline. Theplurality of linear conductive strips are essentially parallel to eachother and oriented substantially along a second direction transverse tothe first direction. In one aspect of this embodiment, the coplanarstripline conductors and the plurality of linear conductive strips arearranged with respect to each other so as to realize a quality factorenhancement and a phase velocity reduction of components of a voltagestanding wave on the coplanar stripline conductors.

Another embodiment of the invention is directed to an SWO employing atapered coplanar stripline configuration so as to significantly reducepower consumption by the SWO. In one aspect of this embodiment, the SWOis configured such that a coplanar stripline region of low conductanceper unit length (low G) is positioned at or near points where maximumvoltage amplitudes are expected, so as to reduce power dissipation tothe substrate. Additionally, in another aspect, a coplanar striplineregion of low resistance per unit length (low R) is positioned at ornear points where maximum currents are expected, so as to reduce powerdissipation from the transmission line itself (i.e., series losses).

Another embodiment of the invention is directed to a coplanar striplineSWO employing one or more of a tailored distributed amplificationscheme, a plurality of linear conductive strips disposed in proximity tothe coplanar stripline, and a tapered coplanar stripline configurationso as to implement mode control and reduce overall power dissipation ofthe oscillator.

In yet another embodiment, an SWO is configured with frequencyadjustability that is again optimized to reduce power dissipation whilefacilitating significant adjustments of oscillator frequency. Forexample, one embodiment of the invention is directed to a method forcontrolling a frequency of at least one voltage standing wave on acoplanar stripline, comprising an act of placing at least one frequencycontrol device along the coplanar stripline at a position that isapproximately at a midpoint between a maximum amplitude of the at leastone voltage standing wave and a zero voltage node of the at least onevoltage standing wave.

Another embodiment of the invention is directed to a closed loop (e.g.,circular) SWO based on a ring resonator coplanar striplineimplementation. In particular, the SWO of this embodiment comprises aclosed-loop coplanar stripline including two conductors, and at leastone amplifier disposed between the two conductors at a first location.The two conductors are connected together at a second location differentfrom the first location to provide a zero voltage node for the at leastone voltage standing wave. In various aspects of this embodiment, one ormore of a tailored distributed amplification scheme, a plurality oflinear conductive strips disposed in proximity to the coplanarstripline, and a tapered coplanar stripline configuration may be usedwith the closed loop structure. In another aspect, a particularamplifier configuration involving cross-coupling of the coplanarstripline conductors is employed to facilitate single mode operation,using a particular resonator topology so as to avoid inducingsignificant loss in the oscillator.

It should be appreciated that all combinations of the foregoing conceptsand additional concepts discussed in greater detail below arecontemplated as being part of the inventive subject matter disclosedherein. In particular, all combinations of claimed subject matterappearing at the end of this disclosure are contemplated as being partof the inventive subject matter disclosed herein.

BRIEF DESCRIPTION OF THE DRAWINGS

The accompanying drawings are not intended to be drawn to scale. In thedrawings, each identical or nearly identical component that isillustrated in various figures is represented by a like numeral. Forpurposes of clarity, not every component may be labeled in everydrawing. In the drawings:

FIG. 1 illustrates various examples of conventional transmission lines;

FIGS. 2A and 2B show different views of a conventional coplanarwaveguide (CPW);

FIGS. 3A and 3B show different views of a conventional coplanarstripline (CPS);

FIG. 4A shows a “single-ended” model of distributed line parameters forthe coplanar waveguide of FIGS. 2A and 2B;

FIG. 4B shows a “differential” model of distributed line parameters forthe coplanar stripline of FIGS. 3A and 3B;

FIGS. 5A and 5B are perspective and cross-sectional views, respectively,illustrating an example of an apparatus based on a coplanar striplineconfiguration, according to one embodiment of the invention;

FIGS. 6A, 6B and 6C are three graphs illustrating a simulated qualityfactor Q vs. signal frequency for different configurations of theapparatus of FIGS. 5A and 5B, according to various embodiments of theinvention;

FIGS. 7A, 7B and 7C are three graphs illustrating a simulated slowingfactor or phase velocity reduction vs. signal frequency for thedifferent configurations represented in FIGS. 6A, 6B, and 6C, accordingto various embodiments of the invention;

FIG. 8 is a cross-sectional view of an exemplary apparatus based on acoplanar stripline configuration, according to yet another embodiment ofthe invention;

FIGS. 9A and 9B are two graphs respectively comparing quality factor Qand slowing factor or phase velocity reduction for different apparatusbased on the configurations shown in FIGS. 5A and 5B, and FIG. 8,according to various embodiments of the invention;

FIG. 10 illustrates a cross-sectional view of an exemplary apparatusbased on a coplanar stripline configuration, according to anotherembodiment of the invention;

FIG. 11 illustrates a perspective view of an exemplary apparatus basedon a coplanar stripline configuration, according to another embodimentof the invention;

FIG. 12 illustrates an example of a conventional standing waveoscillator based on a coplanar stripline implementation;

FIG. 13A illustrates an example of a quarter-wavelength coplanarstripline standing wave oscillator, according to one embodiment of theinvention;

FIG. 13B illustrates voltage and current waveforms for the oscillatorshown in FIG. 13A;

FIG. 14A illustrates an example of a quarter-wavelength coplanarstripline standing wave oscillator employing multiple amplifiers,according to one embodiment of the invention;

FIG. 14B illustrates a voltage waveform for the oscillator shown in FIG.14A;

FIG. 15A illustrates a quarter-wavelength standing wave oscillatoremploying a tapered coplanar stripline configuration according to oneembodiment of the invention;

FIG. 15B is a reproduction of the voltage and current waveforms of FIG.13B, positioned with respect to FIG. 15A so as to illustrate variousconcepts relating to the tapered coplanar stripline configuration,according to one embodiment of the invention;

FIG. 16 illustrates a method according to one embodiment of theinvention for varying R and G along a tapered coplanar stripline withoutaltering the characteristic impedance Z_(o) of the stripline;

FIG. 17 further illustrates the method of FIG. 16 in connection with apiecewise tapered coplanar stripline, according to one embodiment of theinvention;

FIG. 17A illustrates the effects of transistor loading in the exemplaryconfiguration of FIG. 17, according to one embodiment of the invention;

FIG. 17B illustrates a method flow diagram for the design of a piecewisetapered coplanar stripline configuration, according to one embodiment ofthe invention;

FIGS. 18A, 18B and 18C illustrate photographs of three different (λ/4)coplanar stripline standing wave oscillator designs according to variousembodiments of the present invention;

FIGS. 19A and 19B illustrate different representations of frequencyadjustability components for a standing wave oscillator, according toone embodiment of the invention;

FIGS. 20A and 20B illustrate examples of a closed loop standing waveoscillator, according to one embodiment of the invention;

FIG. 21 illustrates an example of a closed loop standing waveoscillator, according to another embodiment of the invention; and

FIGS. 22A and 22B illustrate exemplary signals resulting from asimulation of the closed loop standing wave oscillator of FIG. 21.

DETAILED DESCRIPTION

As discussed above in the Summary, various embodiments of the presentdisclosure are directed to methods and apparatus involving semiconductordevices based on coplanar striplines (CPS). Applicants have recognizedand appreciated that a variety of coplanar stripline configurations mayform the basis of several useful high-speed microelectronic devices fora host of applications. Examples of CPS-based devices incorporatingvarious concepts according to the present disclosure include, but arenot limited to, impedance matching devices, devices for power combiningand division, delays, resonators, oscillators, filters, amplifiers,mixers and the like, including CMOS-based implementations of suchdevices.

In general, high-speed microelectronic devices based on coplanarstripline implementations according to various embodiments of thepresent invention may support differential signals in a range ofapproximately from 1 Gigahertz to approximately 100 Gigahertz, althoughit should be appreciated that the present disclosure is not limited inthese respects. For example, in some implementations based on theconcepts disclosed herein, devices may be configured for operation in avariety of frequency ranges to support either single-ended ordifferential signals.

In the embodiments discussed further below, CPS-based devices mayincorporate various features that dramatically increase the qualityfactor Q of the resulting device. Additionally, an enhancement of thequality factor Q may be achieved while at the same time reducing thephase velocity of one or more waves propagating in the device, therebyalso facilitating the fabrication of relatively smaller devices.

In the section following immediately below, embodiments relating todifferent coplanar stripline configurations according to the presentdisclosure that may be generally employed in a variety of devices arepresented first. Later sections of this disclosure provide some specificexamples of devices based on various coplanar stripline configurations,including particular examples of standing wave oscillators (SWOs). Itshould be appreciated that the examples discussed herein are providedprimarily for the purpose of illustrating some salient conceptsunderlying the present disclosure, and that the invention is not limitedto any particular manner of implementation or any particular examplediscussed herein.

I. Coplanar Striplines with Floating Conductor Arrays

FIGS. 5A and 5B are perspective and cross-sectional views, respectively,illustrating an example of an apparatus 60 based on a coplanar striplineconfiguration, according to one embodiment of the invention. In the topleft corner of FIG. 5A, a coordinate system including an x-axis 36, ay-axis 38 and a z-axis 34 provides a general orientation for theperspective view of the apparatus 60; similarly, in FIG. 5B, the y-axis38 and z-axis 34 in the top left corner indicate that thecross-sectional view is along a direction parallel to the x-axis 36.

As shown in FIG. 5A, the apparatus includes a coplanar stripline 100having a first conductor 100A and a second conductor 100B essentiallyparallel to each other and oriented substantially along a firstdirection parallel to the z-axis 34. The apparatus 60 also includes anarray 62 of essentially linear conductive strips disposed in proximityto the coplanar stripline 100. The linear conductive strips constitutingthe array 62 are essentially parallel to each other, and the array 62 isoriented substantially along a second direction transverse to the firstdirection. In one aspect of this embodiment, as illustrated in FIG. 5A,the second direction may be essentially parallel to the x-axis 36,namely, orthogonal to the first direction along which the coplanarstripline 100 is oriented. It should be appreciated that the number ofconductive strips depicted in the array shown in FIGS. 5A and 5B isprimarily for purposes of illustration, and that the invention is notlimited to any particular number of conductive strips in the array 62.

As also shown in both FIGS. 5A and 5B, the apparatus 60 further includesat least one dielectric material 101 disposed at least between thecoplanar stripline 100 and the array 62 of conductive strips, and asubstrate 103 on which the dielectric material, the array of conductivestrips and the coplanar stripline are disposed. In one aspect of thisembodiment, the dielectric material 101 may be silicon oxide, althoughthe invention is not limited in this respect as other dielectricmaterials may be employed in various implementations. In another aspectof this embodiment, the substrate 103 may be silicon; however, again,the invention is not limited in this respect as other substrates (e.g.,GaAs, SiGe, etc.) may be employed in various implementations. Withreference to FIG. 5B, it may be seen that according to another aspect ofthis embodiment, the coplanar stripline 100 (of which only the conductor100B is visible in the view of FIG. 5B) is disposed in a first plane 64,and the array 62 of linear conductive strips is disposed in a secondplane 66 essentially parallel to the first plane 64, such that a normal65 to both the first plane and the second plane passes through both ofone conductor of the coplanar stripline and at least one conductivestrip of the array 62.

According to other aspects of the apparatus 60 of the embodiment shownin FIGS. 5A and 5B, the apparatus is configured generally to support atleast one signal on the coplanar stripline having a frequency in a rangeof from approximately 1 Gigahertz (GHz) to approximately 100 GHz. Morespecifically, the apparatus may be configured to support a signal on thecoplanar stripline having a frequency in a range of from approximately10 GHz to 60 GHz. In various implementations, differential signals (orsingle-ended signals) may be transported along the conductors 100A and100B of the coplanar stripline while the array of linear conductivestrips is maintained at a floating electric potential with respect tothe conductors 100A and 100B. As discussed further below, the proximityof the floating conductor array 62 to the coplanar stripline 100 resultsin a dramatic increase in the quality factor Q of the apparatus,relative to that generally observed in a conventional coplanar striplinewithout the array 62 (e.g., refer to FIGS. 3A and 3B).

For example, in one aspect of the embodiment of FIGS. 5A and 5B, thecoplanar stripline 100 and the array 62 of linear conductive strips arearranged such that the apparatus has a quality factor Q of at least 30for at least one frequency in the range of from approximately 1 GHz toat least 60 GHz. In another aspect, the coplanar stripline and the arrayof linear conductive strips are arranged such that the apparatus has aquality factor Q of at least 50 for at least one frequency in the rangeof from approximately 1 GHz to at least 60 GHz. In yet another aspect,the coplanar stripline and the array of linear conductive strips arearranged such that the apparatus has a quality factor Q of at least 70for at least one frequency in the range of from approximately 1 GHz toat least 60 GHz. As discussed further below, the foregoingcharacteristics are achieved at least in part via the selection ofparticular dimensions of the various components of the apparatus,particular spacings between the components, and the types of materialsemployed in the apparatus.

According to yet another aspect of this embodiment, the presence of thefloating conductor array 62 in the apparatus 60 of FIGS. 5A and 5B mayalso result in a reduction of the phase velocity of one or more wavespropagating in the device, thereby also facilitating the fabrication ofrelatively smaller devices. This “slow-wave” effect of the floatingconductor array is known in connection with other structures, in which aperiodic loading of a waveguide or transmission line by such floatingconductors generally is considered to spatially separate the electricand magnetic energy in the propagating wave(s). Such a separation ofelectric and magnetic energy results in an increased capacitance perunit length C of the structure. According to the relationshipν=1/√{square root over (LC)}, such an increased capacitance per unitlength C in turn results in a smaller phase velocity ν, and hence asmaller wavelength λ at a given signal frequency f.

To facilitate a discussion of both Q-enhancement and phase velocityreduction effects in the apparatus 60, a number of dimensions forvarious components and spacings between components are indicated inFIGS. 5A and 5B, as well as some physical characteristics (e.g.,permittivity ε and conductivity σ) for materials employed in theapparatus.

For example, with respect to the coplanar stripline 100, a width 68 ofeach of the first and second conductors 100A and 100B along a directionparallel to the x-axis 36 is indicated in FIG. 5A with the notation W.Similarly, a space 70 or distance between the first and secondconductors is indicated with the notation S. Accordingly, a dimension 72for the overall width of the coplanar stripline 100 is indicated in FIG.5A with the notation D, wherein D=2W+S. A thickness 74 for each of theconductors 100A and 100B, along a direction parallel to the y-axis 38,is indicated in both FIGS. 5A and 5B with the notation t_(cps). Finally,an overall length 96 of the coplanar stripline 100 along a directionparallel to the z-axis 34 in the apparatus 60 is indicated in both FIGS.5A and 5B with the notation L_(CPS).

With respect to the array 62 of linear conductive strips, a length 76 ofeach strip, along a direction parallel to the x-axis 36, is denoted inFIG. 5A with the notation l_(s). Similarly, a width 78 of each stripalong a direction parallel to the z-axis 34, shown in both FIGS. 5A and5B, is denoted as d_(A), whereas a space 80 along this direction,between neighboring strips of the array (also shown in both FIGS. 5A and5B), is denoted as d_(B). A thickness 84 along a direction parallel tothe y-axis for each of the strips of the array 62 is denoted in bothFIGS. 5A and 5B as t_(strip), whereas a distance 82 along this directionbetween the first plane 64 (in which lies the coplanar stripline 100)and the second plane 66 (in which lies the array 62) is denoted asd_(S).

With respect to the dielectric material 101 and the substrate 103 of theapparatus 60 shown in FIGS. 5A and 5B, a dielectric thickness ordistance 86 along a direction parallel to the y-axis, between the secondplane 66 and an upper boundary of the substrate 103, is denoted in bothFIGS. 5A and 5B by d_(die), and a permittivity 90 of the dielectricmaterial is denoted by ε_(die). Similarly, a substrate thickness ordistance 88 along a direction parallel to the y-axis is denoted asd_(sub), a permittivity 92 of the substrate is denoted as ε_(sub), and aconductivity 94 of the substrate is denoted as σ_(sub).

In general, as discussed above, Applicants have recognized andappreciated that the selection of particular dimensions of the variouscomponents of the apparatus 60, particular spacings between thecomponents, and the types of materials employed in the apparatus notonly determine the range of frequencies for which the apparatus iscapable of effectively carrying signals, but also determine the degreeof Q-enhancement and phase velocity reduction realized in the apparatus.In particular, through both simulation and empirical processes, a numberof useful generalizations have been established regarding the overallconfiguration of the apparatus 60 and, more specifically, the lengthl_(s), width d_(A) and spacing d_(B) of the conductive strips of thearray 62, with respect to one or both of Q-enhancement and phasevelocity reduction.

For example, according to one aspect of the embodiment of FIGS. 5A and5B, in general, favorable conditions for significant Q-enhancement mayinclude configurations in which the length l_(s) of the conductivestrips of the array 62 and the overall width D of the coplanar stripline100 are approximately equal. More specifically, in one aspect, dramaticQ-enhancement may be observed in structures in which the length l_(s) ofthe conductive strips is slightly greater (e.g., up to approximately 10%greater) than the overall width D of the coplanar stripline.

In other aspects, favorable conditions for significant Q-enhancementalso may include configurations in which at least one of the width d_(A)of each conductive strip and the space d_(B) between neighboringconductive strips is significantly less than the overall width D of thecoplanar stripline. More specifically, favorable Q-enhancementconfigurations may include those in which one or more of the followingconditions is found: the width d_(A) and the space d_(B) are at leastone order of magnitude less than the overall width D of the coplanarstripline; the width d_(A) of the conductive lines and the space d_(B)between conductive lines are approximately one order of magnitude lessthan the overall width D of the coplanar stripline; and the width d_(A)and the space d_(B) are approximately equal.

According to yet other aspects, favorable Q-enhancement configurationsmay include those in which one or more of the following conditions isfound: at least one of the width d_(A) of each conductive strip and thespace d_(B) between neighboring conductive strips is significantly lessthan the overall length L_(CPS) of the coplanar stripline; the widthd_(A) and the space d_(B) are at least one order of magnitude less thanthe overall length L_(CPS) of the coplanar stripline; the width d_(A) ofthe conductive lines and the space d_(B) between conductive lines areapproximately one order of magnitude less than the overall lengthL_(CPS) of the coplanar stripline; and the width d_(A) and the spaced_(B) are approximately equal.

FIGS. 6A, 6B and 6C are three graphs illustrating a number of plots ofsimulated quality factor Q (vertical axis of the graphs) vs. signalfrequency in GHz (horizontal axis of the graphs) for the apparatus 60shown in FIGS. 5A and 5B using a variety of different values for thevarious dimensions and spacings discussed above (i.e., the length l_(s)of the conductive strips of the array 62, the width d_(A) of eachconductive strip, and the space d_(B) between neighboring conductivestrips). It should be appreciated that the particular structuressimulated to provide the graphs of FIGS. 6A, 6B and 6C are merelyexemplary, and that various apparatus pursuant to the present disclosureare not limited to the simulated examples. The simulated devices and theresults generated therefrom are discussed herein primarily for purposesof illustrating some of the concepts discussed immediately above withrespect to exemplary favorable conditions for Q-enhancement.

In the simulations reflected in the graphs of FIGS. 6A, 6B and 6C, thesubstrate 103 of the apparatus 60 shown in FIGS. 5A and 5B is siliconhaving a thickness d_(sub) of 250 micrometers, a permittivity ε_(sub) of11.9, and a conductivity σ_(sub) of 10 Siemens/meter. The dielectricmaterial 101 is silicon oxide having a thickness d_(die) of 5.155micrometers and a permittivity ε_(die) of 4.0. The width W of each ofthe conductors 100A and 100B of the coplanar stripline 100 is 80micrometers, and the space S between the conductors is 60 microns, suchthat the overall width D of the coplanar stripline is 220 micrometers.The thickness t_(cps) of each conductor 100A and 100B is 0.925micrometers, the space d_(s) between the coplanar stripline and thearray 62 is 1.0 micrometer, and the thickness t_(strip) of eachconductive strip is 0.64 micrometers. Finally, the length L_(CPS) of thesimulated apparatus is 400 micrometers.

With the foregoing values used as constants for all simulationsresulting in the graphs of FIGS. 6A, 6B, and 6C, the length l_(s) of theconductive strips of the array 62, the width d_(A) of each conductivestrip, and the space d_(B) between neighboring conductive strips wereeach varied independently of the other to observe the effect on thequality factor Q of the apparatus. Table 1below summarizes the variousvalues for these parameters used in the simulations resulting in thegraphs of FIGS. 6A, 6B and 6C, and is followed by a more detaileddiscussion of the graphs. Each plot referenced in Table 1 and indicatedin FIGS. 6A, 6B, and 6C represents a different simulated apparatus.TABLE Length l_(s) Width d_(A) Space d_(B) Plot No. (microns) (microns)(microns) 150 240 5 5 152 400 5 5 154 180 5 5 156 240 5 5 158 240 1 5160 240 10 5 162 240 20 5 164 240 5 5 166 240 5 10 168 240 5 20 170 2405 0.5

FIG. 6A shows three plots 150, 152 and 154 respectively representingapparatus having three different lengths l_(s) for the conductivestrips, while the width d_(A) and the space d_(B) both are held constantat 5 micrometers each. In particular, the plot 150 reflects a lengthl_(s) of 200 micrometers (slightly longer than the width D of thecoplanar stripline), the plot 152 reflects a length l_(s) of 400micrometers (significantly longer than the width D of the coplanarstripline), and the plot 154 reflects a length l_(s) of 180 micrometers(less than the width D of the coplanar stripline).

From the graph of FIG. 6A, it may be readily observed that at afrequency near 30 GHz, the highest quality factor Q of approximately 65is obtained in the simulated apparatus in which the length l_(s) isapproximately equal to and slightly longer than the width D of thecoplanar stripline. As discussed further below in connection with FIG.9A, it should be appreciated however that each of the simulated devicesin FIG. 6A achieves significant Q-enhancement relative to an apparatusbased on a coplanar stripline using similar dimensions and materials,but without the array 62 of conductive metal strips. In particular, thequality factor Q of such devices without the array 62 remains less than10 in the frequency range of from approximately 5-60 GHz (see plot 176of FIG. 9A). Hence, the addition of the array 62 to such an apparatus(e.g., as shown in FIGS. 5A and 5B) generally results in significantQ-enhancement throughout this frequency range for a variety of differentlengths l_(s) of the conductive strips.

FIG. 6B shows four plots 156, 158, 160 and 162 respectively representingapparatus having four different widths d_(A) for the conductive strips,while the length l_(s) for the strips is held constant at 240micrometers and the space d_(B) between the strips is held constant at 5micrometers. In particular, the plot 156 reflects a width d_(A) of 5micrometers (i.e., equal to the space d_(B)); thus, this plot isidentical to the plot 150 shown in FIG. 6A. The plot 158 in FIG. 6Breflects a width d_(A) of 1 micrometer (significantly less than thespace d_(B)), the plot 160 reflects a width d_(A) of 10 micrometers(twice the space d_(B)) and the plot 162 reflects a width d_(A) of 20micrometers (significantly more than the space d_(B)).

From the graph of FIG. 6B, it may be readily observed that at afrequency near 30 GHz, the highest quality factor Q of approximately 65is obtained in the simulated apparatus in which the width d_(A) and thespace d_(B) both are 5 micrometers, which is significantly less than theoverall width D and length L_(CPS) of the coplanar stripline 100. Again,though, it should be appreciated that each of the simulated devices inFIG. 6B, with the exception of the device represented by the plot 162(in which the width d_(A) is significantly more than the space d_(B)),achieves significant Q-enhancement (Q>10) relative to an apparatus basedon a coplanar stripline using similar dimensions and materials, butwithout the array 62 of conductive strips (e.g., see plot 176 of FIG.9A). In the case of the plot 162 of FIG. 6B, the significantly greaterwidth d_(A) relative to space d_(B) may cause the conductive strips tobegin to resemble a conductive plate below the coplanar stripline ratherthan an array, thereby undermining the role of the array 62 towardreducing losses in the apparatus and enhancing the quality factor Q.

FIG. 6C shows four plots 164, 166, 168 and 170 respectively representingapparatus having four different spaces d_(B) between neighboringconductive strips, while the length l_(s) for the strips is heldconstant at 240 micrometers and the width d_(A) of each strip is heldconstant at 5 micrometers. In particular, the plot 164 reflects a spaced_(B) of 5 micrometers (i.e., equal to the width d_(A)), the plot 166reflects a space d_(B) of 10 micrometers (twice the width d_(A)), theplot 168 reflects a space d_(B) of 20 micrometers (significantly morethan the width d_(A)) and the plot 170 reflects a space d_(B) of 0.5micrometers (significantly less than the width d_(A)).

It should be appreciated that the plot 164 of FIG. 6C is identical tothe plot 156 of FIG. 6B and the plot 150 of FIG. 6A, namely, both thewidth d_(A) and the space d_(B) are 5 micrometers, for which the highestQ of 65 was obtained in the simulations of FIGS. 6A and 6B at afrequency near 30 GHz. From the graph of FIG. 6C, however, it isinteresting to note from the plot 166 that at a frequency near 30 GHz, aslightly higher quality factor Q of approximately 70 is obtained in thesimulated apparatus in which the space d_(B) is 10 micrometers and thewidth d_(A) is 5 microns. Also noteworthy from plot 166 is that thehighest Q of 75 for the simulations in FIG. 6C is obtained with thisconfiguration at a frequency of about 35 GHz.

In any case, in the simulations of FIG. 6C, again both of the dimensionsfor the width d_(A) and space d_(B) are significantly less than theoverall width D and length L_(CPS) of the coplanar stripline 100. Also,each of the simulated devices in FIG. 6C achieves significantQ-enhancement (Q>10) relative to an apparatus based on a coplanarstripline using similar dimensions and materials, but without the array62 of conductive strips (e.g., see plot 176 of FIG. 9A). In the case ofthe plot 170 of FIG. 6C, the somewhat less dramatic Q-enhancementresulting from the significantly greater width d_(A) (5 micrometers)relative to space d_(B) (0.5 micrometers) again may be due to theconductive strips beginning to resemble a conductive plate below thecoplanar stripline rather than an array, thereby undermining the role ofthe array 62 toward reducing losses in the apparatus and enhancing thequality factor Q.

FIGS. 7A, 7B and 7C are three graphs illustrating plots of a “slowingfactor” or phase velocity reduction (vertical axis of the graphs) vs.signal frequency in GHz (horizontal axis of the graphs) respectivelycorresponding to the simulations illustrated in the graphs of FIGS. 6A,6B, and 6C. In particular, the plots 150′, 152′ and 154′ of FIG. 7Acorrespond to the same simulation conditions (see Table 1) as the plots150, 152 and 154 of FIG. 6A, while the plots of FIGS. 7B and 7C have asimilar correspondence with the plots of FIGS. 6B and 6C. In the graphsof FIGS. 7A, 7B and 7C, and as discussed elsewhere herein, the “slowingfactor” is defined as c/ν, wherein c represents the wave velocity in air(i.e., c=1/√{square root over (ε_(o)μ_(o))}), and ν represents the phasevelocity in the given simulated coplanar stripline-based apparatus.

As can be readily observed in the graphs of FIGS. 7A, 7B and 7C, all ofthe simulated apparatus based on the dimensions given in Table 1 exhibitsome significant degree of phase velocity reduction. It is interestingto note, however, that the plots indicating the greatest degree of phasevelocity reduction in the graphs of FIGS. 7A, 7B and 7C (i.e., plots152′, 156′ and 170′) do not necessarily correspond in all cases to theplots indicating the greatest degree of Q-enhancement in the graphs ofFIGS. 6A, 6B and 6C (e.g., compare the plot 150 of FIG. 6A to the plot150′ of FIG. 7A). Hence, these graphs illustrate the appreciable degreeof latitude in designing various CPS-based apparatus according to thepresent disclosure, and “optimizing” an apparatus for a particularapplication. Stated differently, specific dimensions for variouscomponents of an apparatus according to the present disclosure may beselected at least in part based on the respective importance in a givenapplication of size reduction (which relates to phase velocityreduction) and loss (which relates to the quality factor Q).

Again, it should be appreciated that the particular structures simulatedto provide the graphs of FIGS. 6A, 6B, 6C, 7A, 7B and 7C are merelyexemplary, and that various apparatus pursuant to the present disclosureare not limited to the specific materials and dimensions employed inthese examples. In sum, however, these simulations generally demonstratethat both significant Q-enhancement and phase velocity reduction may berealized in coplanar stripline-based apparatus according to variousembodiments of present invention. These simulations also providenoteworthy guidelines for configurations of such apparatus in whichQ-enhancement and phase velocity reduction may be observed.

FIG. 8 is a diagram illustrating a cross-sectional view (similar to thatof FIG. 5B) of an apparatus 60A according to yet another embodiment ofthe invention. In FIG. 8, the apparatus 60A includes two arrays 62A and62B of essentially linear conductive strips, wherein one of the arrays62A is disposed in the second plane 66, and another of the arrays 62B isdisposed in a third plane 67 essentially parallel to the first plane 64and the second plane 66. According to one aspect of the embodiment shownin FIG. 8, the conductive strips of the arrays 62A and 62B are arrangedin an alternating fashion, such that no normal to the first, second andthird planes passes through both a conductive strip of the array 62A anda conductive strip of the array 62B. The multiple arrays 62A and 62Bemployed in the apparatus 60A of FIG. 8 generally facilitate a furtherphase velocity reduction as compared to the apparatus 60 shown in FIGS.5A and 5B, while at the same time maintaining an appreciable degree ofQ-enhancement as compared to a coplanar stripline-based apparatuswithout any array(s) of conductive strips.

FIGS. 9A and 9B illustrate two graphs, of quality factor Q vs. frequencyand slowing factor or phase velocity reduction vs. frequency,respectively, comparing results of simulations based on themultiple-array apparatus 60A of FIG. 8, the single-array apparatus 60 ofFIGS. 5A and 5B, and a similarly-dimensioned coplanar striplineapparatus without any array(s) of conductive strips (e.g., see FIGS. 3Aand 3B). Specifically, in FIG. 9A, the plot 172 represents simulationresults of Q vs. frequency for the single-array apparatus 60, the plot174 represents simulation results of Q vs. frequency for themultiple-array apparatus 60A, and the plot 176 represents simulationresults of Q vs. frequency for a coplanar stripline apparatus withoutany array(s) of conductive strips. In FIG. 9B, the plot 172′ representssimulation results of slowing factor vs. frequency for the single-arrayapparatus 60, the plot 174′ represents simulation results of slowingfactor vs. frequency for the multiple-array apparatus 60A, and the plot176′ represents simulation results of slowing factor vs. frequency for acoplanar stripline apparatus without any array(s) of conductive strips.

In the graphs of FIGS. 9A and 9B, again a silicon substrate and asilicon oxide dielectric material are employed in all of the simulatedapparatus, with material parameters (ε_(die), ε_(sub), σ_(sub)) and asubstrate thickness d_(sub) identical to those discussed above inconnection with the simulations represented in FIGS. 6A, 6B and 6C.Additionally, the coplanar stripline dimensions W, S, D, L_(CPS) andt_(CPS) are identical to those discussed above in connection with FIGS.6A, 6B and 6C. For the single-array and multiple-array apparatussimulations of FIGS. 9A and 9B, the length l_(s) of each conductivestrip is 240 micrometers, the width d_(A) of each strip is 5micrometers, the space d_(B) between neighboring strips of the samearray is 5 micrometers, and the thickness t_(strip) of each conductivestrip is 0.64 micrometers. For the multiple-array apparatus, withreference to FIG. 8, the distance d_(s) between the first and secondplanes, as well as the second and third planes, is 1.0 micrometer, andthe dielectric thickness d_(die), between the third plane 67 and theboundary with the substrate 103, is 3.515 micrometers.

As can be readily observed in FIG. 9A, while the multiple-arrayapparatus, as represented by the plot 174, does not achieve as high aquality factor Q as the single-array apparatus (represented by the plot172), both the multiple-array and single-array apparatus achieve somesignificant degree of Q-enhancement as compared to a coplanarstripline-based apparatus without any array(s), as represented by theplot 176. More specifically, the plot 176 (representing essentially acoplanar stripline on a silicon substrate) remains significantly below aQ of 10 for most of the frequency range between approximately 5 GHz and60 GHz, whereas the plots 172 and 174 remain significantly above a Q of10 for most of this frequency range.

In FIG. 9B, it is readily observed that the multiple-array apparatus, asrepresented by the plot 174′, achieves a significantly higher slowingfactor or phase velocity reduction than the single-array apparatusrepresented by the plot 172′. However, again both the single-arrayapparatus and the multiple-array apparatus achieve significant phasevelocity reduction as compared to the coplanar stripline-based apparatuswithout any array(s), represented by the plot 176′.

In yet other embodiments, different numbers and arrangements of multiplearrays of conductive strips may be employed together with a coplanarstripline to facilitate one or both of Q-enhancement and phase velocityreduction. For example, FIG. 10 illustrates a cross-sectional view(similar to that of FIGS. 5B and 8) of an apparatus 60B, according toone embodiment of the invention, which employs three arrays 62A, 62B and62C of essentially linear conductive strips. The apparatus 60B of FIG.10 is substantially similar to that shown in FIG. 8, with the exceptionof the addition in FIG. 10 of the array 62C disposed in a fourth plane69 parallel to the first plane 64, the second plane 66 and the thirdplane 67. FIG. 11 illustrates a perspective view (similar to that ofFIG. 5A) of yet another apparatus 60C, according to one embodiment ofthe invention, which employs two arrays 62A and 62D of conductivestrips, wherein the array 62A and 62D are disposed respectively belowand above the coplanar stripline 100. According to different aspects ofthe embodiment of FIG. 11, the individual conductive strips of one ofthe arrays 62A and 62D may be arranged in vertical alignment (i.e.,along the y-axis) with a corresponding strip of another of the arrays62A and 62D or, alternatively, the individual strips of the arrays maybe arranged in an alternating manner (e.g., similar to themultiple-array arrangements shown in FIGS. 8 and 10). In still otherembodiments, one or more arrays of conductive strips may be arrangedabove and/or below the coplanar stripline 100 in a variety of manners.

II. Coplanar Stripline Standing Wave Oscillators

Having discussed various concepts relating to coplanar striplineconfigurations according to the present disclosure that may be generallyemployed in a variety of different devices, exemplary coplanar striplinedevices according to other embodiments of the present invention, basedon a standing wave oscillator, are now presented. It should beappreciated that, according to different embodiments discussed in detailbelow, standing wave oscillators according to the present invention mayor may not be configured with one or more arrays of conductive strips,as discussed above in Section 1.

a. Background

One of the most basic and ubiquitous building blocks of communicationssystems, as well as a host of other applications, is the oscillator.Essentially all communications systems at some point need a referenceoscillator to facilitate a variety of communication-related functions.As a result, oscillator design in the high frequency regime is an activearea of interest. In particular, electromagnetic wave effects that needto be accounted for as system frequency dramatically increases have ledto significant interest in various high frequency oscillator designsbased on transmission lines.

Various types of oscillators based on transmission line implementationsconventionally have been employed to generate high frequency clocksignals in the gigahertz (GHz) range. Many of these conventionalapproaches ultimately are directed at generating essentially square waveclock signals that may be globally distributed throughout an integratedcircuit system without significant propagation-delay-induced phaseshifts. More specifically, these approaches generally aim to generate aglobal clock signal with low clock skew and low clock jitter that may bepropagated across an entire system in a manner that preserves thecorrect ordering of events throughout the system. Both traveling waveoscillators (TWOs) and standing wave oscillators (SWOs) based ontransmission line implementations have been employed for such purposes.

Standing waves are of particular interest in connection with the presentdisclosure due to the unique characteristics of such waves. A standingwave is formed when two waves traveling in opposite directions withidentical amplitude and frequency interact. Unlike a traveling wave,which has an amplitude and phase that varies over time at a givenposition along a transmission line, a standing wave has a constantamplitude and phase at a given position along a transmission line,wherein the amplitude varies sinusoidally with position along the line.A common way to generate a voltage standing wave is to send an incidentwave down a transmission line and reflect it back with a losslesstermination such as a short circuit. However, losses from thetransmission line conductors themselves (e.g., “series” losses due to Rand “parallel” losses due to G) typically cause amplitude mismatchbetween the incident and reflected waves, resulting in a residualtraveling wave which distorts the standing wave. Accordingly, toeffectively implement a self-sustaining standing wave oscillator, sometype of compensation scheme (i.e., amplification) must be employed toovercome the losses inherent in the transmission line.

One conventional implementation of a standing wave oscillator using acoplanar stripline is shown in FIG. 12. In FIG. 12, a coplanar stripline100 (similar to that shown in FIGS. 3A and 3B) having conductors 100Aand 100B is configured to form a half-wave (λ/2) resonator 200 byshorting out both ends of the length of the coplanar stripline, therebyforcing a voltage standing wave “node” (i.e., zero potential between theconductors 100A and 100B) at both ends of the coplanar stripline.Theoretically, the resonator 200 supports at least one standing wavehaving a frequency related to λ, wherein the amplitude of the wavevaries along the length of the resonator, as shown schematically at thebottom of FIG. 12.

In the oscillator configuration of FIG. 12, the effect of coplanarstripline conductor loss on signal amplitude is offset by providingdistributed amplifiers (i.e., transconductors) spaced along the lengthof the resonator to provide a distributed transconductance. Inparticular, FIG. 12 illustrates a number of NMOS cross-coupled-pairtransconductors 104A, 104B and 104C supplied by three respective currentsources 106A, 106B and 106C. Each of these transconductors is coupled tothe conductors 100A and 100B of the coplanar stripline 100 at adifferent position along the coplanar stripline. A number of PMOSdiode-connected loads 108A, 108B and 108C also are coupled to thecoplanar stripline 100 to establish a common-mode voltage between theconductors 100A and 100B.

It is noteworthy that in the configuration of FIG. 12, thetransconductors 104A, 104B and 104C are configured to have a same gain.The gain of a given transconductor is related to the current provided bythe current source associated with the transconductor (one of thecurrent sources 106A, 106B and 106C) multiplied by the width of thetransistors making up the transconductor (i.e., transistor gainincreases with increasing width and/or current). In the oscillatorconfiguration shown in FIG. 12, each transistor of a cross-coupled pairhas a same width, and each transconductor is provided with a samecurrent; hence, the transconductors all have a same gain. The use ofmultiple same gain transconductors to compensate for conductor loss onthe coplanar stripline permits an equivalent lumped model for such acoplanar stripline that facilitates a relatively straightforwarddetermination of the resonator parameters required to supportoscillation at a given frequency.

However, one issue that arises from employing multiple transconductorshaving a same gain in the configuration of FIG. 12 is that significantenergy arguably is wasted due to over-amplification. More specifically,again with reference to the wave illustration at the bottom of FIG. 12,it should be readily appreciated that for the wave mode illustrated, theamplitude of the wave near the center of the resonator structure has amaximum value and steadily decreases away from the center and towardseither end of the resonator. Hence, to support the illustrated mode, thetransconductors 104A and 104C, which are configured to have the samegain as the transconductor 104B located at the center of the resonator,arguably are configured for a higher amplification than necessary; inparticular, these transconductors utilize more current than necessary,thereby wasting valuable power resources.

Yet another issue that arises from the oscillator configuration of FIG.12 is that the resonator does not employ any mode control mechanism(e.g., suppression of higher order modes). As a result, thisconfiguration has the tendency to excite strong high frequency modes.The lack of mode control in this configuration ultimately deterioratesthe quality of the generated sinusoidal signal, as the presence ofmultiple higher frequency modes distorts the sinusoidal waveform at thefundamental resonance frequency.

For example, the equal-gain amplifiers evenly distributed along thelength of the resonator structure shown in FIG. 12 theoretically cansupport a mode at λ/2, as shown in FIG. 12, as well as other oddharmonics such as λ, (3/2)λ, (5/2)λ, 3λ, etc.; specifically, eachamplifier is capable of acting as an electrically open node that cansupport a higher frequency mode. In this manner, the resonator of FIG.12 is not optimized for sinusoidal waveform generation. This conditionmay be significantly undesirable for many applications. It should beappreciated, however, that since the ultimate use of the resonator shownin FIG. 12 is for an essentially square-wave clock generator, thepresence of some higher order modes may not substantially affect theoverall performance of the resonator toward generating such a clocksignal.

Applicants have recognized and appreciated that conventional standingwave oscillators (SWOs) based on coplanar stripline implementations maybe modified and improved to facilitate the generation of high quality,high frequency sinusoidal signals. The general frequency range forsinusoidal signals generated by SWOs considered by the presentdisclosure includes frequencies from approximately 1 GHz to 100 GHz,although it should be appreciated that the present disclosure is notlimited in this respect. According to various embodiments of the presentinvention discussed further below, single mode SWOs may be particularlyconfigured to generate sinusoidal signals in these exemplary frequencyranges with low power dissipation and low phase noise. A number ofcontemplated applications for such oscillators include, but are notlimited to, communications systems including wireless communications,radar, phase-locked loops (PLL) for various applications, and the like.

b. Quarter-Wavelength Coplanar Stripline Standing Wave Oscillators

FIGS. 13A and 13B illustrate some fundamental concepts underlying acoplanar stripline SWO according to one embodiment of the invention. Inparticular, FIG. 13A illustrates an essentially quarter-wavelength (λ/4)coplanar stripline SWO 300 that includes conductors 300A and 300Bforming a differential coplanar stripline (i.e., similar to the coplanarstripline 100 of FIGS. 3A and 3B). The SWO 300 is formed by a length L(reference numeral 301) of the coplanar stripline, with one end of thecoplanar stripline terminated by a short 302, and the other end of theline terminated by a pair of cross-coupled inverters serving as anamplifier 304. In another aspect of this embodiment, the amplifier 304may be implemented in a manner similar to that shown in FIG. 12 by anNMOS cross-coupled transconductor pair driven by a current source (forclarity of the drawing, the NMOS transistors and the current source arenot shown explicitly in FIG. 13A, but represented schematically by thecross-coupled inverters). Such an amplifier forms an active positivefeedback network, converting DC energy into RF energy and injecting thisenergy into the circuit to compensate losses relating to the coplanarstripline.

The SWO 300 shown in FIG. 13A is configured to support standing wavesthat satisfy the boundary conditions at both ends of the coplanarstripline, namely, having a maximum voltage amplitude swing at theamplifier end of the coplanar stripline and a zero voltage node at theshorted end of the line. Accordingly, the possible excitation modesbased on a length L of coplanar stripline theoretically correspond toL=λ/4+n(λ/2) (for n=0, 1, 2, 3 . . . ). In practical implementations,the actual length L that supports excitation modes may differ slightlyfrom the theoretical lengths for any of a variety of reasons discussedfurther below. The fundamental frequency of oscillation f_(o)corresponds to n=0, that is f_(o)=ν/(4L), where ν is the wave's phasevelocity as determined by the materials both surrounding andconstituting the coplanar stripline.

FIG. 13B schematically illustrates the voltage and current waves alongthe length of the SWO 300, indicated respectively in the figure as V(z)and I(z), for the fundamental mode supported by the SWO 300. The graphof FIG. 13B is illustrated along a z-axis corresponding to the length ofthe SWO, where z=0 corresponds to the position of the amplifier 304 andz=L corresponds to the position of the shorted end. From FIG. 13B, itmay be readily appreciated that while the voltage amplitude swing V(z)is maximum at the left in the drawing (z=0) and decreases to zero movingto the right and arriving at the shorted end (z=L), the current I(z)behaves in an opposite manner; namely, the current is minimum at theleft and increases moving to the right, being maximum at the shorted endof the coplanar stripline. According to one aspect of this embodiment,an output of the SWO may be derived across the amplifier 304 (i.e., thepoint of maximum voltage amplitude swing), wherein the output isappropriately buffered to reduce any load on the SWO.

c. Standing Wave Oscillators with Distributed/Tailored Gain Cells

Another embodiment of the invention based on the (λ/4) coplanarstripline SWO 300 shown in FIG. 13A is illustrated in FIG. 14A andrelates to distributed amplification. It should be appreciated, however,that the concepts discussed in connection with this embodiment may beimplemented in a variety of other SWO configurations according to thepresent invention, as discussed herein. Accordingly, the particularexample discussed immediately below relating to a quarter-wavelength SWOis provided primarily for purposes of illustration.

To facilitate an explanation of this embodiment, the voltage waveformshown in FIG. 13B is reproduced in FIG. 14B. In the embodiment of FIG.14A, a plurality of amplifiers or “gain cells” 304A, 304B, 304C . . .304D are deployed along the length of the coplanar stripline. While FIG.14A explicitly illustrates four such amplifiers, it should beappreciated that the invention is not limited in this respect, asdifferent numbers of amplifiers may be employed in SWOs according to theinvention. Also, while the amplifiers are schematically represented inFIG. 14A as being equally spaced along the coplanar stripline, theinvention also is not limited in this respect, as a variety of positionsfor the amplifiers are possible according to different embodiments. Ingeneral, it should be appreciated that the number and placement ofamplifiers according to various embodiments of the invention, as well asthe respective gains of the amplifiers, may depend at least in part onone or more desired modes to be excited by the oscillator, as discussedfurther below.

For example, according to one aspect of the embodiment shown in FIG.14A, the relationship between the respective gains of the amplifiers aretailored to approximate the relationship between the expected voltageamplitudes of the desired standing wave mode at different positionsalong the coplanar stripline at which the amplifiers are deployed. Forexample, with reference to FIG. 14B, since the voltage amplitude of theillustrated standing wave mode decreases from left to right along thelength of the coplanar stripline, the respective gains G₁, G₂, G₃ . . .G_(n) of the amplifiers also are decreased from left to right along thecoplanar stripline (i.e., moving from z=0 to z=L). In this manner, thegains of the amplifiers in this example are “amplitude-dependent.”

In the embodiment shown in FIG. 14A, some well-known benefits ofdistributed amplification (e.g., increased frequency response) areachieved while at the same time conserving valuable power resources bytailoring the gains of the amplifiers. Recall that, in a conventionalSWO implementation shown in FIG. 12, multiple distributed amplifiers areconfigured to have same gains notwithstanding the different voltageamplitudes at the amplifier deployment positions; hence, in thisconventional configuration, significant energy arguably is wasted due toover-amplification. In contrast, the SWO implementation of FIG. 14Ausing tailored gain amplifiers according to the present inventionrequires less total current to operate than a similar implementationusing multiple same-gain amplifiers, thereby conserving valuable powerresources.

Moreover, the multiple tailored gain amplifiers of the embodiment shownin FIG. 14A additionally function as a mode control mechanism to ensureoscillation substantially in a single mode (e.g., corresponding to λ/4).Again, this is in contrast to the conventional configuration usingmultiple same-gain amplifiers shown in FIG. 12 which theoretically cansupport a number of other modes, thereby degrading the sinusoidalquality of a signal generated by the oscillator.

To compare and contrast the embodiments of FIGS. 13A (one lumpedamplifier) and FIG. 14A (distributed amplifiers), exemplaryimplementations of these SWOs employing a coplanar stripline having alength of approximately 1500 micrometers have been analyzed. In theexemplary implementation corresponding to FIG. 14A, four amplifiers areused along the coplanar stripline, placed at equal intervals of z=0,z=L/4, z=L/2 and z=3L/4. Recall that, as discussed above, the gain of agiven amplifier is proportional to the square root of the product oftransistor size and current drawn. All of the transistors used in theamplifiers have lengths of 0.18 microns. The transistor widths andcurrent drawn by each amplifier, which determine the amplifier gain,respectively are as follows: Amplifier Position Transistor Widths(microns) Current Drawn (mA) z = 0 22.5 12 z = L/4 22.5[sin(3π/8)]12[sin(3π/8)] z = L/2 22.5[sin(π/4)] 12[sin(π/4)] z = 3L/422.5[sin(π/8)] 12[sin(π/8)]In this implementation, the SWO oscillates at 12.19 GHz, with a maximumvoltage amplitude at z=0 of 2.09 Volts.

In the exemplary implementation corresponding to FIG. 13A, the gain ofthe single amplifier 304 is chosen to be equal to the total lumped gainof the distributed amplifiers used in the implementation correspondingto FIG. 14A. More specifically, the length of the transistors of theamplifier 304 is again 0.18 micrometers, and the transistor widths aregiven by 22.5[1+sin(3λ/8)+sin(7λ/4)+sin(7λ/8)] microns. Similarly, thetotal current conducted by the amplifier is12[1+sin(3π/8)+sin(7λ/4)+sin(7λ/8)] milliamperes. This SWO oscillates at9.76 GHz, with a maximum voltage amplitude at z=0 of 2.27 Volts. Hence,while the exemplary lumped amplifier SWO based on FIG. 13A achieves ahigher amplitude, the exemplary distributed amplifier SWO based on FIG.14A achieves an appreciably higher frequency of operation.

e. Standing Wave Oscillator Employing a Tapered Coplanar Stripline

FIG. 15A illustrates another embodiment of a (λ/4) coplanar striplineSWO 500 according to the invention, wherein the SWO is based on atapered coplanar stripline configuration with position-dependent lineparameters. To facilitate an explanation of the embodiment shown in FIG.15A of the SWO 500 employing a tapered configuration, the voltage andcurrent waveforms shown in FIG. 13B for a (λ/4) coplanar stripline SWOare reproduced in FIG. 15B. It should be appreciated, however, that theconcepts discussed in connection with this embodiment may be implementedin a variety of other SWO configurations according to the presentinvention, as discussed herein. Accordingly, the particular examplediscussed immediately below relating to an essentiallyquarter-wavelength SWO is provided primarily for purposes ofillustration. Additionally, as discussed below, it should be appreciatedthat a tapered coplanar stripline configuration according to the presentinvention is not limited in application for use in a SWO, and may beemployed in other CPS-based devices.

1. Coplanar Striplines with Position-Dependent Parameters

One embodiment of the present invention is directed to a coplanarstripline that is configured such that the resistance per unit length Rand the conductance per unit length G are discrete or continuousfunctions of position along the coplanar stripline (i.e., R(z) andG(z)). In one aspect of this embodiment, the coplanar stripline may befurther configured such that a uniform characteristic impedance issubstantially maintained notwithstanding variations in R and G, so as toavoid local reflections.

In one exemplary implementation of this embodiment, as shown for examplein the SWO 500 illustrated in FIG. 15A, a tapered coplanar striplineconfiguration is employed wherein a space 504 between coplanar striplineconductors 500A and 500B, and/or a width 502 of each conductor 500A and500B, vary discretely or continuously as a function of position z alongthe coplanar stripline. FIG. 15A is a top view of the taperedconfiguration 500 (similar to the view of FIG. 3B), wherein the space504 in FIG. 15A is accordingly indicated with the notation S(z) and thewidth 502 is accordingly indicated with the notation W(z). In otherrespects, the tapered configuration 500 may be similar to that shown inthe cross-sectional view of FIG. 3A; namely, the conductors 500A and500B may be disposed on a dielectric material above a substrate. Thetapered configuration of the conductors 500A and 500B in FIG. 15Aeffectively changes the coplanar stripline parameters R and G along thelength of the coplanar stripline so that they are position dependent,while effectively maintaining a uniform characteristic impedance of thecoplanar stripline.

In particular, the resistance per unit length R generally relates to thewell-known skin effect, wherein at higher frequencies charge carrierstravel closer to the edges and away from the core of a given conductor.As the two conductors making up the coplanar stripline are brought incloser proximity to each other (i.e., as the distance S decreases and/orthe conductor width W increases), the respective charges flowing nearthe edge or “skin” of the conductors are brought more closely together,thereby impeding charge flow. Hence, as the conductors are brought moreclosely together, generally the resistance per unit length R increases.

The conductance per unit length G generally relates to electromagneticfield loss between the conductors and the substrate above which thecoplanar stripline is disposed. With reference again particularly to thecoplanar stripline cross-section illustrated in FIG. 3A, as conductorsof a coplanar stripline are moved farther apart from each other (i.e.,as the distance S increases and/or the conductor width W decreases), thefields due to current flowing through the conductors have more of anopportunity to interact with the substrate above which the coplanarstripline is disposed; hence, the conductance per unit length Gincreases. Conversely, as the conductors are brought more closelytogether (i.e., as the distance S decreases and/or the conductor width Wincreases), the loss to the substrate generally decreases, and hence theconductance per unit length G decreases.

In sum, from the foregoing, it should be appreciated that the coplanarstripline parameters R and G in the above example generally varyinversely with conductor separation; namely, as the conductors arebrought more closely together, R increases and G decreases; conversely,as the conductors are separated by a greater distance, R decreases and Gincreases.

2. Implications of Position-Dependent Parameters for an SWO

With respect to signal propagation on a coplanar stripline in general, Rmay be viewed as coupling to current waves whereas G may be viewed ascoupling to voltage waves to introduce respective series and shuntlosses; accordingly, smaller R corresponds to less series loss, andsmaller G corresponds to less shunt loss. This tradeoff between seriesloss R and shunt loss G, due to their inverse variation with conductorseparation, may impose a major constraint with respect to lossminimization in a coplanar stripline carrying traveling waves. However,when a coplanar stripline hosts a standing wave, as shown in FIG. 15B,the R-G tradeoff may be exploited via the tapered configuration 500shown in FIG. 15A to take advantage of the position-dependent standingwave amplitude so as to significantly reduce loss (and correspondinglyenhance the quality factor Q of the resulting device).

For example, from FIG. 15B it may be appreciated that at z=0, where thevoltage amplitude swing of the SWO 500 of FIG. 15A is maximum, a lowerconductance per unit length G results in less power loss to thesubstrate, as power loss to the substrate is proportional to the squareof the voltage (relatively high at z=0) multiplied by the conductanceper unit length. Accordingly, even with relatively high voltage at thispoint, loss to the substrate can be reduced by having a coplanarstripline configuration with a low conductance per unit length G. On theother hand, at z=0, FIG. 15B illustrates that the current flowing in theconductors of the coplanar stripline is at a minimum; accordingly, anypower loss due to the coplanar stripline conductors (i.e., due to theresistance per unit length R) is of less concern, since this power lossis proportional to the square of the current (relatively low at z=0)multiplied by the resistance per unit length R. Hence, even if R is highat this point, it does not necessarily induce significant losses, due tothe low current.

The opposite scenario holds for z=L (i.e., at the shorted end of thecoplanar stripline shown in FIG. 15A). In particular, as shown in FIG.15B, at this point, the voltage is zero, and the current is at amaximum. Hence, having a significant resistance per unit length R atthis point in the coplanar stripline would result in significant lossdue to the high current, whereas the conductance per unit length G is ofrelatively lesser concern due to the low voltage (i.e., a zero voltagenode).

In view of the foregoing, one embodiment of the invention is directed toa quarter-wavelength SWO including a coplanar stripline with varyingresistance per unit length R(z) and varying conductance per unit lengthG(z), in which a region of low conductance per unit length (low G) ispositioned at the point z=0 where a maximum voltage amplitude isexpected, so as to reduce power dissipation to the substrate.Additionally, the SWO is configured such that a coplanar striplineregion of low resistance per unit length (low R) is positioned at thepoint z=L where a maximum current is expected. The SWO 500 of FIG. 15Aprovides one example of such an arrangement. In general, according tothis embodiment, the position-dependent voltage and current amplitudesresulting from the standing wave facilitate a reduction in device losses(and corresponding Q enhancement) by appropriately tailoring theparameters R and G based on the fixed position amplitudes.

The tapered coplanar stripline configuration employed in this embodiment(as well as other embodiments) may be implemented in a number ofdifferent ways. For example, according to one aspect, the overall lengthof the coplanar stripline may be divided into a discreet number of equalor varying length sections each having a different R and G, where L andC are kept constant to maintain a substantially uniform characteristicimpedance to effectively prevent local reflections. Alternatively, thecoplanar stripline may be implemented with a gradually taperingconductor spacing and width such that R and G vary gradually withposition along the coplanar stripline, while again maintaining asubstantially uniform characteristic impedance.

FIG. 16 includes a graph and a corresponding exemplary tapered coplanarstripline configuration 505 illustrating a method according to oneembodiment of the invention for varying R and G along the striplinewithout significantly altering the characteristic impedance Z_(o) of thestripline. According to one aspect of this embodiment, the graph of FIG.16 may be compiled from data acquired through computer simulations(e.g., Sonnet EM) based on varying the width W of the striplineconductors and the space S between the stripline conductors along thelength of the stripline. Hence, the horizontal axis of the graph in FIG.16 represents the width W and the vertical axis of the graph representsthe space S between conductors of the stripline.

The graph of FIG. 16 includes plots of three exemplary “constantcharacteristic impedance contours” Z_(o,1), Z_(o,2) and Z_(o,3); inparticular, each of these contours represents a different constantcharacteristic impedance for varying values of Wand S, whereinZ_(o,3)>Z_(o,2)>Z_(o,1). FIG. 16 also includes plots of three exemplary“loss contours” (R₁,G₁), (R₂,G₂) and (R₃,G₃), wherein each loss contourreflects a constant value for R and a corresponding constant value for Gfor varying values for W and S. Although the graph of FIG. 16 representseach of the loss contours as a single line representing an identicalconstant value for both R and G, in reality the respective values for Rand G along a given loss contour are not identical, but nonethelessappreciably close to each other. Thus, in the graph of FIG. 16, it is areasonable approximation for practical design purposes to assume thatthe values of R and G for each loss contour are virtually identical.

As illustrated in FIG. 16, increasing either W or S results in adecreased R and increased G due to the R-G tradeoff discussed above(i.e., R₃>R₂>R₁ and G₃<G₂<G₁). However, the characteristic impedanceZ_(o) increases with increasing S but decreases with increasing W.Accordingly, to achieve low G near z=0 and low R at z=L so as to reduceloss, without significantly affecting Z_(o), the coplanar striplineconductors may be simultaneously widened and moved apart from z=0 toz=L, following one of the Z_(o) contours shown in FIG. 16.

To illustrate the foregoing concepts, the design of a tapered coplanarstripline configuration having an essentially constant characteristicimpedance Z_(o,2) from the graph of FIG. 16 is considered as an example.It should be appreciated that the methodology underlying this example,as discussed below, may be applied similarly to other characteristicimpedance contours representing a desired characteristic impedance forthe resulting device.

Specifically, with reference to the constant characteristic impedancecontour Z_(o,2) in FIG. 16, three points A, B, and C are identifiedalong the Z_(o,2) contour, at the respective intersections of thiscontour with the loss contours (R₃, G₃), (R₂, G₂) and (R₁, G₁). As alsoillustrated in the example of FIG. 16, the dimensions W_(A) and S_(A)corresponding to the point A (i.e., high R, low G) are used for theportion of the tapered stripline 505 around z=0, the dimensions W_(B)and S_(B) corresponding to the point B are used for the portion aroundthe middle of the stripline, and the dimensions W_(C) and S_(C)corresponding to the point C (i.e., low R, high G) are used for theportion of stripline around z=L.

While the foregoing example utilizes three points of reference A, B, andC along the characteristic impedance contour Z_(o,2) to determinecorresponding dimensions along the tapered coplanar striplineconfiguration 505, it should be appreciated that the invention is notlimited in this respect; namely, any number of points along a givencharacteristic impedance contour theoretically may be used to determinecorresponding dimensions along the tapered coplanar stripline. Inparticular, as the number of points increases, the resulting taperedcoplanar stripline increasingly resembles one in which R and G areessentially continuous functions of position z along the stripline. Itshould be appreciated, however, that for virtually any finite number ofpoints along a given impedance contour, a piecewise taperedconfiguration results in which R and G vary discretely (i.e., in apiecewise fashion) along the stripline.

FIG. 17 further illustrates the concept of such a piecewise variation.In particular, FIG. 17 includes a graph in W-S space showing a plot ofan exemplary impedance contour representing a constant characteristicimpedance Z_(o) (note that the W-S axes in the graph of FIG. 17 areexchanged from those of FIG. 16). Five points (1, 2, 3, 4, and 5) areselected along this contour, corresponding to respective W and Sdimensions for five different portions or sections of a piecewisetapered coplanar stripline configuration 505, shown directly below thegraph in FIG. 17 (the exemplary dimensions W₅ and S₅ are indicated inFIG. 17 for section 5). Although five points are chosen in the exampleof FIG. 17, again it should be appreciated that different numbers ofpoints may be chosen in other embodiments. As also illustratedqualitatively in FIG. 17, the length along the z-axis of each section1-5 of the piecewise tapered coplanar stripline may or may not be thesame as one or more other sections of the stripline; in particular,according to various embodiments, an optimal apportionment of eachsection 1-5 relative to the overall length of the tapered coplanarstripline may be determined by a mathematical procedure (discussed indetail below), and optionally adjusted by empirical determination.

More specifically, in some embodiments of the piecewise taperedconfigurations illustrated in FIGS. 16 and 17, loss considerations maydictate the particular respective lengths and positions of each sectionin a piecewise configuration. For example, in one embodiment, tominimize the overall loss of the tapered configuration, each sectioncould be placed at a given position z which would yield the minimumlocal loss at z, given the standing wave voltage and current amplitudesat that position.

However, since the standing wave amplitudes in the z-domain (i.e., V(z)and I(z)) are dependent upon the tapered coplanar stripline structureitself (and hence unknown before the construction of the stripline) thedesign and construction of a loss-optimized tapered stripline viewedfrom the perspective of the z-domain generally may be somewhatchallenging, and require a time-consuming and perhaps costly iterativeapproach. In view of the foregoing, Applicants have recognized andappreciated that the design and construction of a tapered striplineconfiguration may be significantly facilitated by considering the designfrom the perspective of the θ-domain, where θ is the wave's phase.

In particular, as discussed in detail below, the standing wave voltageand current amplitudes in the θ-domain may be considered for practicalpurposes to be simple sinusoids (assuming weak loss); accordingly,applying a transformation from the z-domain to the θ-domainsignificantly simplifies the loss analysis for the design. After designof a piecewise tapered configuration in the θ-domain, an inversetransformation may be applied to render design parameters in thez-domain, which is necessary to obtain the actual dimensions (i.e.,section lengths along the z-axis) for the physical layout of the taperedconfiguration. In the following discussion, this process is detailedstep by step.

The overall time-averaged loss, P_(diss), in a general tapered(position-dependent) coplanar stripline with a constant characteristicimpedance, which hosts a single standing wave mode, is given by$\begin{matrix}{P_{diss} = {\int_{0}^{L}{\left\{ {{\frac{1}{2}{R(z)}{I^{2}(z)}} + {\frac{1}{2}{G(z)}{V^{2}(z)}}} \right\}{\mathbb{d}z}}}} & (1)\end{matrix}$where L is the horizontal span of the line, I(z) and V(z) are thecurrent and voltage amplitudes of the standing wave mode at position z,and R(z) and G(z) are the series resistance and shunt conductance perunit length at z. In order to obtain the minimum-loss tapered line, oneneeds to find R(z) and G(z) that minimize P_(diss) in Eq. (1) under theconstraint of the R-G trade-off discussed above. However, it is verydifficult to evaluate the integration in Eq. (1), since I(z) and V(z)are not known a priori, as they depend on the physical structure of thestripline, which has yet to be determined. Accordingly, the designprocess in the z-domain poses somewhat of a circular argument; inparticular, a time-consuming iterative approach would be required,making the optimization procedure very involved and potentially costly.

According to one embodiment of the invention, evaluation of Eq. (1)becomes substantially simplified by a transformation in which theintegration variable z is replaced by θ, the wave's phase. First,consider a piecewise tapered configuration having an infinitesimalnumber of uniform segments. Each uniform segment of the piecewiseconfiguration has a length dz and the identical characteristic impedanceZ_(o). Traveling down the infinitesimal uniform line segment locatedbetween z and z+dz, a wave experiences an infinitesimal phase change ofdθ, where dθ and dz are related through dθ=β(z)dz. Here β(z) is thepropagation constant of the traveling wave in the infinitesimal uniformsegment and is given by the familiar formulaβ(z)=ω/ν(z)=ω√{square root over (L(z)C(z))},  (2)where ν(z)=1/√{square root over (L(z)C(z))} is the wave's phase velocityin the infinitesimal uniform line segment, L(z) and C(z) are inductanceand capacitance per unit length in the infinitesimal uniform segment,and ω is the modal frequency. Substituting in the above relationβ(z)=dθ/dz, we obtain the following relation between θ and z:dθ=ω√{square root over (L(z)C(z))}dz, or  (3)θ(z)=ω∫₀ ^(z)√{square root over (L(z′)C(z′))}dz′  (4)Again, in the case of a uniform line, θ(z) reduces to the familiarω√{square root over (LC)}z=βz, where β is the phase constant, 2π/λ. Butin a non-uniform line, the wave phase velocity ν(z)=1/√{square root over(L(z)C(z))} may vary with z, so θ(z) is not a linear function.

Mapping from z to θ(z) is useful because in any general transmissionline with a constant characteristic impedance Z_(o) the voltage andcurrent amplitudes for a standing wave mode are always sinusoids of thephase θ(z), assuming weak loss. Accordingly, these amplitudes may bere-written as:V(z)=V ₀ cos(θ(z))  (5)I(z)=I ₀ sin(θ(z)).  (6)With the parameterization to θ, the power dissipation equation from Eq.(1) may be re-written asP _(diss)=∫₀ ^(π/2){½(I ₀ sin θ)² R _(θ)(θ)+½(V ₀ cos θ)² G(θ)}dθ  (7)assuming the line length is chosen so as to produce π/2 phase shift (foran essentially quarter-wavelength SWO). Here R_(θ)(θ) and G_(θ)(θ) aredefined as series and shunt loss per radian phase shift at θ, and arerelated to R(z) and G(z) byR _(θ)(θ)dθ=R(z)dz  (8)G _(θ)(θ)dθ=G(z)dz  (9)where the relationship between dz and dθ may be obtained from Eqs. (3)or (4). The integration in Eq. (7) is relatively easy since the currentand voltage standing waveforms are always known sinusoids in theθ-domain, irrespective of the particular tapered striplineconfiguration.

In view of the foregoing, a particular example of a piecewiseconfiguration based on the concepts discussed above in connection withFIG. 17 may be used to illustrate an optimization process for a designusing the z-domain to θ-domain transformation, according to oneembodiment of the invention. With reference again to the characteristicimpedance contour shown in FIG. 17, loss parameters were simulated forfive points along the contour (points 1-5), based on an exemplarycharacteristic impedance of Z_(o)=25 ohms. Table 2 below provides theresults of this simulation, showing the relevant W-S dimensions of thestripline for each section as well as the corresponding loss parametersR_(θ) and G_(θ). TABLE 2 Point on Z_(o) contour W (μn) S (μn)R_(θ)(mΩ/deg) G_(θ)(μS/deg) 1 75 20 12.4 3.23 2 80 30 9.72 4.00 3 85 506.16 7.96 4 90 100 3.26 19.0 5 90 120 1.96 25.3

Once the loss parameters in the θ-domain are obtained for each of thefive sections, the amount of phase change that each section shouldcontribute to minimize the overall loss of the tapered configuration maybe determined. According to one embodiment, this can be done byevaluating, at each point in the θ-domain (i.e., 0≦θ≦π/2), which of thefive sections minimizes the loss per unit phase shift at that localpoint. The loss per unit phase shift is the integrand of the lossintegral in Eq. (7): $\begin{matrix}{\frac{\mathbb{d}P_{diss}}{\mathbb{d}\theta} = {{\frac{1}{2}\left( {I_{o}\sin\quad\theta} \right)^{2}{R_{\theta}(\theta)}} + {\frac{1}{2}\left( {V_{o}\cos\quad\theta} \right)^{2}{{G_{\theta}(\theta)}.}}}} & (10)\end{matrix}$

With reference to FIG. 17, for purposes of illustrating the z-θtransformation, the z-axis is also labeled as the θ-axis, and transitionpoints (θ₁, z₁), (θ₂, z₂), (θ₃, z₃), and (θ₄, z₄) are indicated at theboundaries between the sections. The transition points θ₁, θ₂, θ₃, andθ₄ between the sections may be calculated with the aid of Eq. (10) byequating the loss per unit phase shift of one section to that of thenext section. For example, θ₁ can be calculated by${{\frac{1}{2}\left( {I_{o}\sin\quad\theta_{1}} \right)^{2}R_{\theta,1}} + {\frac{1}{2}\left( {I_{o}Z_{o}\cos\quad{\theta\quad}_{1}} \right)^{2}G_{\theta,1}}} = {{\frac{1}{2}\left( {I_{o}\sin\quad\theta_{1}} \right)^{2}R_{\theta,2}} + {\frac{1}{2}\left( {I_{o}Z_{o}\cos\quad\theta_{1}} \right)^{2}G_{\theta,2}}}$where R_(θ,1) and R_(θ,2) are the series resistance per unit phase shiftfor sections 1 and 2, respectively (from Table 2), while G_(θ,1) andG_(θ,2) are the shunt conductance per unit phase shift for sections 1and 2, respectively (again from Table 2). This calculation yieldsθ₁=22.9° for the particular example given in Table 2. Therefore, section1 has lower loss per unit phase shift than section 2 for θ<θ₁=22.9° andsection 2 has lower loss per unit phase shift than section 1 forθ>θ₁=22.9°. Thus, in one exemplary design, section 1 should span roughlythe first 22.9° of the tapered stripline configuration, and at the 22.90point, there should be a transition to section 2. The phase spans andcorresponding transition points θ₂, θ₃, and θ₄ of the other sections maybe determined similarly; for example, applying Eq. (10) as above for θ₂,θ₂ is found to be 39.8°, so the phase span of section 2 is approximately17° (i.e., θ₂-θ₁).

Having obtained the transition points between each section (and hencethe span of each section) in the θ-domain according to Eq. (10), thesevalues are then transformed to the z-domain to yield the correspondingtransition points z₁, z₂, z₃ and z₄ (see FIG. 17) and therefore therespective physical lengths of the different sections of the piecewisedesign. To this end, with reference to FIG. 17, the physical length ofthe i-th section (i=1, 2, 3, 4, 5) is given by Δz_(i)=z_(i)−z_(i−1) inthe z-domain, which corresponds to the phase span Δθ_(i)=θ_(i)−θ_(i−1)in the θ-domain. Using Eq. (3) above, these two quantities are relatedbyΔθ_(i)=ω√{square root over (L _(i) C _(i))}Δz_(i)  (11)where L_(i) and C_(i) are the inductance and capacitance per unit lengthfor the i-th section, and are known from the EM simulations.Accordingly, Eq. (11) may be used to determine the length of eachsection in the z-domain, completing the transformation of the designfrom the θ-domain to the z-domain.

According to another aspect of this embodiment, as an optional furtherstep in the process outlined above, once the physical length of eachsection is determined according to the above procedure the actualstripline layout near the transition points z₁, z₂, z₃ and z₄ may besmoothed so that the line further approximates or becomes an essentiallycontinuous tapered configuration. The values of W and S thus becomeinterpolations of the original selected points that were simulated. Asmentioned above, it should be appreciated that the more points/sectionsone selects and simulates for the piecewise design, the better optimizedthese interpolated values become.

At this point, for SWO design using a tapered stripline, the piecewisetapered design may be schematically simulated to determine anyadjustment that may be required in the design to account for theboundary condition associated with the amplifier 304 shown in FIG. 15A.The transistors of the amplifier effectively introduce additional phaseshift to that of the coplanar stripline itself. Therefore, if the SWO issimulated using a coplanar stripline spanning a quarter of thewavelength corresponding to the target frequency, the actual oscillationfrequency may be lower than this target.

Accordingly, in one aspect of this embodiment, to compensate for theloading effects of the amplifier the stripline configuration may beshortened until the simulated oscillation frequency reaches the targetfrequency. For example, if the target oscillation frequency is 20 GHz,and if the simulated oscillation frequency does not reach 20 GHz until15° worth of phase shift is removed from the stripline, then this 15°may be eliminated from the beginning of the line in the layout. In theparticular example discussed above in connection with FIG. 17 and Table2, the phase span parameter Δθ₁ of section 1 may be shortened from 22.9°down to 7.9° to account for the effects of amplifier loading. Thismodification is illustrated in FIG. 17A, in which a portion 507 ofsection 1 indicated in shading and crossed-out with an “X” is removedfrom the stripline.

In sum, it should be appreciated that a design procedure for a piecewisetapered coplanar stripline configuration according to one embodiment ofthe invention, as outlined above in connection with the specificexamples given in FIG. 17 and Table 2, is provided primarily forpurposes of illustration, and that the disclosure is not limited to thisexample. In particular, the salient concepts underlying this designprocedure may be generally specified as follows, with reference to themethod flow diagram illustrated in FIG. 17B: 1) select a characteristicimpedance Z_(o) for the piecewise tapered configuration; 2) select thenumber of sections to be included in the piecewise tapered configuration(i.e., select the number of points in a graph of contours similar tothose shown in FIGS. 16 and 17); 3) for each section, determine in theO-domain the loss parameters R_(θ) and G_(θ) based on Eqs. (8) and (9);4) determine in the O-domain the transition points between sectionsbased on Eq. (10); and 5) transform the transition points (or phasespans) in the θ-domain to the z-domain, based on Eq. (11), to determinethe respective physical lengths of the different sections. As anoptional additional step, once the physical length of each section isdetermined, the transition points may be smoothed via interpolation ofthe width W and spacing S. As a further option, for SWO designs based onpiecewise tapered configurations, amplifier loading effects may becompensated by shortening the overall length of the stripline (e.g., asillustrated in FIG. 17A).

It should be appreciated that although the exemplary tapered coplanarstripline configurations illustrated in FIGS. 15A, 16, 17 and 17A arebased on an essentially (λ/4) coplanar stripline SWO, the invention isnot limited in this respect. In particular, tapered transmission lineconfigurations having a variety of dimensional profiles may beimplemented for different types of devices in which different R and/or Gvalues may be desirable at different points along the device. Ingeneral, tapered transmission line configurations according to variousembodiments of the invention may be designed to have arbitrary values ofR and/or G as a function of position z along the transmission line for avariety of applications.

f (λ/4) Coplanar Stripline SWO with Q-Enhancement and Phase VelocityReduction Features

FIGS. 18A, 18B and 18C illustrate photographs of three different (λ/4)coplanar stripline standing wave oscillator designs according to variousembodiments of the present invention. In particular, FIG. 18A shows atop view of a circuit die of a uniform coplanar stripline SWO 510 (basedat least in part on the embodiment shown in FIG. 13A), whereas FIGS. 18Band 18C show respective top views of circuit dies of different taperedcoplanar stripline SWOs 512 and 514 (based at least in part on theembodiment shown in FIG. 15A). In each of these (λ/4) coplanar striplineSWOs, the short 302 between the conductors of the stripline at theposition z=L is illustrated in the top portion of the figures, whereasconnection points 516 at the position z=0 for one or more amplifiers(similar to the amplifiers 304 shown in FIGS. 13A and 15A) are indicatedin the bottom portion of the figures.

Each of the SWOs shown in FIGS. 18A, 18B and 18C was fabricated using a0.18 micrometer CMOS technology, and in cross-section each of the SWOsalso includes one or more arrays 62 of conductive strips, similar tothose discussed above in connection with FIGS. 5A, 5B, 8, 10 and 11 (inFIGS. 18A, 18B and 18C, which are top views, the arrays 62 are indicatedgenerally as a shaded area underneath the conductors of the coplanarstriplines). As discussed above in connection with these earlierfigures, in one aspect of these embodiments, the presence of thearray(s) of conductive strips facilitate both Q-enhancement and phasevelocity reduction in the SWOs. In another aspect, loss reductionrealized by the tapered configurations of FIGS. 18B and 18C contributesto further Q-enhancement in these embodiments.

In each of the SWOs shown in FIGS. 18A, 18B and 18C, the significantconductor mass at the short 302 tends to increase series resistance at apoint in the structure where a relatively lower R is desirable.Accordingly, in one embodiment, each of the SWOs may further include aconductive metal plate 63 on a same plane as one or more of the arrays62 (e.g., underneath the short 302, as indicated by the solid white areain the figure), wherein the short 302 is connected to the plate 63 by anumber of vias. This arrangement essentially augments the mass of theconductor in the area of the short 302 and thereby reduces the seriesresistance in this area.

In the tapered embodiment of FIG. 18B, the greater conductor separationat z=L as compared to the uniform embodiment of FIG. 18A results in aproportionally longer short 302. This longer short 302 also may tend toincrease series resistance relative to the structure shown in FIG. 18A,thereby potentially undermining in part the benefits of the taperedconfiguration. In view of the foregoing, the embodiment of FIG. 18Coffers an alternative tapered configuration, in which the taper of thestripline conductors is modified such that the length of the short 302is similar to that of the uniform configuration shown in FIG. 18A.

To comparatively measure the performance of the uniform and taperedconfigurations, the SWOs illustrated in FIGS. 18A, 18B and 18C werefabricated to each have a characteristic impedance Z_(o) ofapproximately 25 ohms for operation at approximately 15 GHz. Each of thedevices has an overall stripline length L of approximately 420micrometers. For the uniform embodiment of FIG. 18A, the width of eachconductor of the stripline is approximately 85 micrometers and thespacing between the conductors is approximately 50 micrometers. For thetapered configuration of FIG. 18B, conductor width ranged fromapproximately 75 micrometers near z=0 to approximately 90 micrometersnear z=L, and the spacing between conductors ranged from approximately20 micrometers at z=0 to approximately 120 micrometers at z=L (e.g., seeTable 2). Experimental measurements confirmed that a Q-enhancement ofapproximately 50% was realized in the tapered device of FIG. 18Brelative to the uniform device of FIG. 18A (e.g., the uniform device hada quality factor Q of approximately 39, whereas the tapered device had aQ of approximately 59).

g. Low Loss Frequency Tunable Standing Wave Oscillators

In yet another embodiment of the present invention, a coplanar striplineimplementation of an SWO may be configured with frequency adjustabilitythat may be again optimized to reduce loss and hence power consumption.For example, according to one embodiment, an SWO is implemented with oneor more variable capacitors (“varactors”) that vary the capacitance perunit length C of the coplanar stripline, and hence vary the frequency ofoscillation (phase velocity ν, which relates frequency and wavelength,is inversely proportional to the square root of the product LC). In oneaspect of this embodiment, the placement of one or more varactors on thecoplanar stripline is optimized to maintain appreciable frequencyadjustability while reducing any losses incurred by the varactor(s).

FIGS. 19A and 19B show different representations of a varactor that maybe employed with an SWO according to one embodiment of the invention. Inparticular, FIG. 19A shows a varactor 400 connected between twoconductors 300A and 300B of a coplanar stripline, wherein the varactoris implemented as a pair of NMOS transistors having their gates coupledto respective conductors of the coplanar stripline, and having theirsources and drains coupled together and connected to a bias voltageVbias. FIG. 19B shows another equivalent schematic representation of thevaractor 400, wherein a variable capacitance 400A is shownseries-connected to a resistance 400B, representing the inherent lossassociated with the varactor 400.

With reference again to FIGS. 13A and 13B illustrating an exemplary(λ/4) coplanar stripline SWO, it should be appreciated that implementingone or more varactors 400 in a SWO according to various embodiments ofthe invention may affect power consumption due to losses associated withthe varactor resistance 400B. In particular, if a varactor is positionedin an SWO at a point of maximum voltage amplitude swing (e.g., z=0 inFIG. 13A), the frequency adjustability is significant, but also lossesdue to a relatively high voltage across the varactor resistance may beappreciable. On the other hand, placing a varactor close to the shortedend of the SWO (e.g., z=L in FIG. 13A) would result in low losses due tolittle or no voltage across the varactor resistance, but also little orno frequency tuning ability.

However, Applicants have recognized and appreciated that, in connectionwith at least some fabrication processes, while loss due to the varactorresistance decreases essentially linearly moving from a point of maximumvoltage amplitude (i.e., z=0) to a voltage node (i.e., z=L), the same isnot true of frequency tuning ability; namely, frequency tuning abilitybased on varactor position remains essentially constant from a point ofmaximum voltage amplitude up to approximately halfway toward a voltagenode (i.e., 0<z≦L/2). After the halfway point (L/2<z≦L), frequencytuning ability begins to noticeably drop off as the voltage node isapproached, at which point there is no frequency tuning ability. Hence,in some processes, while it has been noted that there is an essentiallylinear relationship between varactor position along the resonator andloss due to varactor resistance, there is also a significantlynon-linear relationship between varactor position along the resonatorand frequency tuning ability.

In view of the foregoing, according to one embodiment of the invention,this phenomenon is exploited in a coplanar stripline SWO by positioninga varactor in proximity of a halfway point (e.g., z≈L/2 in FIG. 13A)between a maximum voltage amplitude and a voltage node (zero volts). Inone aspect of this embodiment, the varactor position may be optimized byplacing the varactor in proximity of the halfway point but between thehalfway point and the voltage node (e.g., L/2<z<<L in FIG. 13A). In thismanner, significant frequency tuning ability is maintained whilesignificantly reducing losses attributed to the varactor resistance. Invarious implementations, a varactor as described above may be employedin connection with uniform or non-uniform (e.g., tapered) coplanarstripline configurations, as well as SWO configurations other than the(λ/4) coplanar stripline SWOs discussed herein. In yet otherembodiments, a distribution of varactors along the coplanar striplinecan be used to provide frequency tunability while mitigating anypotential effects relating to losses due to lumped varactor loading.

h. Closed Loop Standing Wave Oscillators

Another embodiment of the invention is directed to a closed loop (e.g.,circular) standing wave oscillator based on a ring resonator coplanarstripline implementation. In one aspect of this embodiment, as discussedin greater detail below, a cross-coupled amplifier configuration isemployed to facilitate single mode operation, using a particularresonator topology so as to avoid inducing significant loss in theoscillator.

More specifically, FIG. 20A illustrates a closed-loop SWO 700 accordingto one embodiment of the present invention, schematically represented asa circular loop. The SWO 700 employs at least two amplifiers 702A and702B (i.e., two pairs of cross-coupled inverters) which counterbalanceloss in the circuit, and a closed loop coplanar stripline 704 (includingconductors 704A and 704B) having an overall path length L on whichstanding waves are formed to meet the boundary condition V(φ)=V(φ+2π),where φ is an arbitrary reference angle from a given reference radius rof the ring structure. The boundary condition results in possible energymodes at L=2πr=nλ (for n=1, 2, 3 . . . ), where r is the radius of thering. The fundamental frequency of oscillation f_(o) corresponding ton=1 then is given by ν/L, where ν is the phase velocity.

The interconnection of the amplifiers 702A and 702B of the SWO 700 shownin FIG. 20A effectively implements a mode control technique for theoscillator. In particular, by connecting point T1 to point B2 and pointT2 to point B1, the ports T1-T2 and B1-B2 are ensured to be in oppositephase (180°), thereby suppressing all even mode harmonics. This evennode suppression makes port L1-L2 always remain “quiet,” i.e., a zerovoltage node. Port R1-R2 is forced also to be a zero voltage node bytapping the power supply for the amplifiers into this port as a commonmode voltage.

FIG. 20B illustrates one example of a physical layout for the ringresonator schematically represented in FIG. 20A. In the layout of FIG.20B, the interconnections between the amplifiers 702A and 702B forimplementing the even mode suppression are positioned proximate eachother so as to introduce negligible time delay as compared to theintentional delay in the ring coplanar stripline. In particular, theshape of the ring coplanar stripline is distorted, while otherwisekeeping its topology in tact, so that port T1-T2 and port B1-B2 arephysically close to each other to reduce interconnection loss betweenthe ports. FIG. 21 illustrates yet another layout for a closed loop SWOpursuant to the concept of FIG. 20B, in the shape of a “clover leaf”such that the amplifiers 702A and 702B are again positioned proximateeach other. In one aspect of the embodiment shown in FIG. 21, four λ/4sections of the coplanar stripline are coupled together to form thecomplete loop.

FIGS. 22A and 22B illustrate simulation results for a 10 GHz closed loopSWO implemented using a Silicon-Germanium (Si—Ge) process withtransistors whose f_(T) is approximately 50 GHz. As illustrated in FIG.22A, each of the “loud ports” (e.g., T1-T2 and B1-B2 in FIG. 20A) has adifferential voltage swing of 1.2 Volts when the oscillator drawsapproximately 5 mA of DC current from a 1.5V power supply. As shown inFIG. 22B, after some initial ringing, the “quiet port” (e.g., L1-L2 inFIG. 20A) stays quiet, as expected.

According to various aspects of this embodiment, a number of conceptsdiscussed above in connection with quarter-wavelength SWOimplementations also may be employed to realize a variety of closed loopcoplanar stripline SWO configurations. For example, in various aspectsof this embodiment, one or both of a tailored distributed amplificationscheme and a variable parameter coplanar stripline configuration (e.g.,tapered coplanar stripline) may be employed with the closed loopstructure. In other aspects, one or both of a tapered coplanar striplineconfiguration (i.e., with position dependent R and G) and array(s) ofconductive strips may be employed to facilitate Q-enhancement and phasevelocity reduction. In yet another aspect, low loss frequency tuningcapability may be implemented in such an SWO using one or moreappropriately positioned varactors.

III. Conclusion

Having thus described several illustrative embodiments, it is to beappreciated that various alterations, modifications, and improvementswill readily occur to those skilled in the art. Such alterations,modifications, and improvements are intended to be part of thisdisclosure, and are intended to be within the spirit and scope of thisdisclosure. While some examples presented herein involve specificcombinations of functions or structural elements, it should beunderstood that those functions and elements may be combined in otherways according to the present invention to accomplish the same ordifferent objectives. In particular, acts, elements, and featuresdiscussed in connection with one embodiment are not intended to beexcluded from similar or other roles in other embodiments. Accordingly,the foregoing description and attached drawings are by way of exampleonly, and are not intended to be limiting.

1-28. (canceled)
 29. A standing wave oscillator to generate at least onevoltage standing wave, comprising: a closed-loop coplanar striplineincluding two conductors; and at least one amplifier disposed betweenthe two conductors at a first location.
 30. The oscillator of claim 29,wherein the at least one amplifier includes means for distributing again of the at least one amplifier in a varying manner along thecoplanar stripline.
 31. The oscillator of claim 29, further comprisingmeans for controlling an oscillation mode of the oscillator.
 32. Theoscillator of claim 29, further comprising means for controlling anoscillation frequency of the oscillator.
 33. The oscillator of claim 53,wherein the at least one amplifier includes at least a first amplifierlocated at the first location, the first location being one-quarter adistance around the closed loop coplanar stripline in a first directionfrom the second location, and a second amplifier located at a thirdlocation one-quarter the distance around the closed loop coplanarstripline in a second direction from the second location, such that thefirst and second amplifiers are opposite each other in the closed loopcoplanar stripline.
 34. The oscillator of claim 33, wherein: a firstconductor of the two conductors at the first location is connected to asecond conductor of the two conductors at the third location; and thesecond conductor at the first location is connected to the firstconductor at the third location.
 35. The oscillator of claim 34, whereinthe closed-loop coplanar stripline is shaped such that the firstlocation is physically proximate to the third location.
 36. Theoscillator of claim 29, wherein the two conductors include a firstconductor and a second conductor, and wherein the coplanar stripline isconfigured to have a resistance per unit length R and a conductance perunit length G that vary along a length of the coplanar stripline. 37.The oscillator of claim 36, wherein the coplanar stripline is configuredto have a substantially uniform characteristic impedance along thelength of the coplanar stripline.
 38. The oscillator of claim 37,wherein the coplanar stripline is configured as a plurality of sections,wherein each section of the plurality of sections has a differentresistance per unit length R and a different conductance per unit lengthG.
 39. The oscillator of claim 37, wherein the coplanar stripline isconfigured such that the resistance per unit length R and theconductance per unit length G vary substantially continuously along thelength of the coplanar stripline.
 40. The oscillator of claim 37,wherein a space between the first and second conductors and a width ofthe conductors is varied along the length of the coplanar stripline. 41.(canceled)
 42. The oscillator of claim 30, wherein the means fordistributing comprises: means for distributing the amplification alongthe coplanar stripline such that the distributed amplification isrelated to an amplitude of the at least one voltage standing wave. 43.The oscillator of claim 29, wherein the at least one amplifiercomprises: a plurality of amplifiers along the coplanar stripline, atleast two amplifiers of the plurality of amplifiers having differentgains.
 44. The oscillator of claim 43, wherein a gain of each amplifierof the plurality of amplifiers relates to an amplitude of the at leastone voltage standing wave at a position at which the amplifier isdisposed along the coplanar stripline.
 45. The oscillator of claim 44,wherein the plurality of amplifiers are equally spaced along the lengthof the coplanar stripline.
 46. (canceled)
 47. The oscillator of claim29, wherein the at least one amplifier is positioned along the coplanarstripline so as to excite at least one desired oscillation mode of theat least one voltage standing wave.
 48. The oscillator of claim 47,wherein the at least one amplifier comprises a plurality of amplifiersat different locations along the coplanar stripline so as to excite theat least one desired oscillation mode of the at least one voltagestanding wave.
 49. The oscillator of claim 48, wherein the plurality ofamplifiers are distributed along the coplanar stripline such that adistributed amplification of the plurality of amplifiers is related toan amplitude of a desired oscillation mode of the at least one voltagestanding wave.
 50. The oscillator of claim 29, further comprising: atleast one frequency control device along the coplanar stripline placedat a position that is approximately at a midpoint between a maximumamplitude of the at least one voltage standing wave and a zero voltagenode of the at least one voltage standing wave.
 51. The oscillator ofclaim 50, wherein the at least one frequency control device is placedalong the coplanar stripline at a position that is between the midpointand the zero voltage node.
 52. The oscillator of claim 51, wherein theat least one frequency control device is placed closer to the midpointthan to the zero voltage node.
 53. The oscillator of claim 29, whereinthe two conductors are connected together at a second location differentfrom the first location to provide a zero voltage node for the at leastone voltage standing wave.